Sub-1V Curvature Compensated Bandgap Reference

Sub-1V Curvature Compensated Bandgap Reference

Master Thesis Performed in Electronic Devices

By

Kevin Tom

Reg. Nr.: LiTH-ISY-EX-3592-2004

Linköping University, 2004.

Sub-1V Curvature Compensated Bandgap Reference

Master Thesis

Electronic Devices

Department of Electrical Engineering

Linköping University, Sweden.

Kevin Tom

Reg. Nr.: LiTH-ISY-EX-3592-2004

Supervisor: Atila Alvandpour

Examiner: Atila Alvandpour

Linköping, Nov. 1, 2004

Avdelning, Institution Division, Department Institutionen för systemteknik 581 83 LINKÖPING Datum Date 2004-11-01 Språk

Language Rapporttyp Report category ISBN Svenska/Swedish

X Engelska/English Licentiatavhandling X Examensarbete ISRN LITH-ISY-EX-3592-2004

C-uppsats

D-uppsats Serietitel och serienummer _{Title of series, numbering} ISSN

Övrig rapport

____

URL för elektronisk version

http://www.ep.liu.se/exjobb/isy/2004/3592/ Titel

Title Kompensering av Andra Ordningens fel i en sub-1V Bandgaps Referens Sub-1V Curvature Compensated Bandgap Reference

Författare

Author Kevin Tom

Sammanfattning Abstract

This thesis investigates the possibility of realizing bandgap reference crcuits for processes having sub-1V supply voltage. With the scaling of gate oxide thickness supply voltage is getting reduced. But the threshold voltage of transistors is not getting scaled at the same rate as that of the supply voltage. This makes it difficult to incorporate conventional designs of bandgap reference circuits to processes having near to 1V supply voltage. In the first part of the thesis a comprehensive study on existing low voltage bandgap reference circuits is done. Using these ideas a power, low-voltage bandgap reference circuit is designed in the second part of the thesis work.

The proposed bandgap reference circuit is capable of generating a reference voltage of 0.730V. The circuit is implemented in 0.18µm standard CMOS technology and operates with 0.9V supply voltage, consuming 5µA current. The circuit achieves 7 ppm/K of temperature coefficient with supply voltage range from 0.9 to 1.5V and temperature range from 0 to 60C.

Nyckelord Keyword

Abstract

This thesis investigates the possibility of realizing bandgap reference circuits for processes having sub 1V supply voltage. With the scaling of gate oxide thickness supply voltage is getting reduced. But the threshold voltage of transistors is not getting scaled at the same rate as that of the supply voltage. This makes it difficult to incorporate conventional designs of bandgap reference circuits to processes having near to 1V supply voltage. In the first part of the thesis a comprehensive study on existing low voltage bandgap reference circuits is done. Using these ideas a low-power, low-voltage bandgap reference circuit is designed in the second part of the thesis work.

The proposed bandgap circuit is capable of generating a reference voltage of 0.730V. The circuit is implemented in 0.18µm standard CMOS technology and operates with 0.9V supply voltage, consuming 5µA current. The circuit achieves 7 ppm/oK of temperature coefficient with supply voltage range from 0.9 to 1.5V and temperature range from 0 to 60o C.

Acknowledgements

First I would like to thank my supervisor and examiner Professor Atila Alvandpour for giving the opportunity to do this project work and giving me valuable guidance and for the helpful discussions during the thesis work.

I am grateful to all members of Electronic Devices Group, Linköping University for their encouragement and support throughout my thesis work.

I would like to thank my class mate Ameya Bhide for the helpful discussions during the thesis period.

Last, but not the least, I would like to express my deep gratitude to my parents for their encouragement and support to my studies in Sweden.

Table of Contents

ABSTRACT ... 1 ACKNOWLEDGEMENTS ... 3 TABLE OF FIGURES ... 7 LIST OF TABLES... 9 1 INTRODUCTION... 11

1.1 ZENER BASED VOLTAGE REFERENCES... 11

1.2 BANDGAP VOLTAGE REFERENCE... 12

1.3 ADVANTAGES OF BANDGAP REFERENCE CIRCUIT... 12

2 TERMINOLOGY AND DEFINITIONS ... 13

2.1 BANDGAP VOLTAGE... 13

2.2 PTATVOLTAGE... 13

2.3 CTATVOLTAGE... 13

2.4 BANDGAP REFERENCE CIRCUIT (BGR)... 13

2.5 VBEVOLTAGE... 13

2.6 PARTS PER MILLION (PPM)... 14

3 BANDGAP VOLTAGE REFERENCE PRINCIPLE... 15

3.1 DERIVATION OF THE TEMPERATURE INDEPENDENT VOLTAGE... 17

3.2 CMOSBANDGAP REFERENCE CIRCUITS... 20

3.3 SOLVING THE OFFSET ERROR... 22

4 CASE STUDY OF LOW VOLTAGE CMOS BANDGAP CIRCUITS ... 23

4.1 DYNAMIC THRESHOLD MOS(DTMOS)TRANSISTORS... 25

4.2 CMOSBGRUSING RESISTIVE SUB-DIVISION... 27

4.3 BGRUSING TRANSIMPEDANCE AMPLIFIER... 29

4.4 BGRUSING DEPLETION TRANSISTORS... 30

4.5 DISADVANTAGES OF BGRCIRCUIT... 32

4.6 BGRCIRCUIT USING BULK BIASING... 33

4.7 THRESHOLD VOLTAGE BASED VOLTAGE REFERENCE... 34

5 PROPOSED LOW VOLTAGE LOW POWER BGR... 35

5.1 BANDGAP REFERENCE CIRCUIT... 36

5.2 WORKING... 36

5.2.1 PTAT Voltage Generation ... 37

5.2.2 Operational Amplifier... 41

5.2.3 Power Supply Independent Biasing ... 43

5.2.4 Startup Circuit ... 45

5.2.5 Curvature Compensation... 47

5.3 DESIGN CHOICES AND SIMULATED RESULTS... 49

5.4 CONCLUSION... 53

Table of Figures

FIGURE 1HYPOTHETICAL BANDGAP REFERENCE CIRCUIT... 16

FIGURE 2CONVENTIONAL BIPOLAR BANDGAP REFERENCE CIRCUIT... 16

FIGURE 3VARIATION OF VREF WITH TEMPERATURE... 19

FIGURE 4 A)VERTICAL NPN TRANSISTOR B)VERTICAL PNP TRANSISTOR... 20

FIGURE 5TYPICAL CMOSBANDGAP REFERENCE CIRCUIT... 21

FIGURE 6OFFSET ERROR COMPENSATION USING CHOPPED OPERATIONAL AMPLIFIER... 22

FIGURE 7CURRENT SUMMING BGR ... 23

FIGURE 8VOLTAGE SUMMING BGR ... 24

FIGURE 9 A)VREF VS.VDD B)VREF VS.TEMPERATURE... 25

FIGURE 10DTMOS CROSS SECTION... 26

FIGURE 11DTMOSBGRCIRCUIT... 26

FIGURE 12 DTMOSBGRTEMPERATURE DEPENDENCE... 27

FIGURE 13 BGR BASED ON RESISTIVE SUB DIVISION... 27

FIGURE 14VREF VARIATION IN [11]... 28

FIGURE 15BGRCIRCUIT WITH IMPROVED NOISE IMMUNITY... 29

FIGURE 16BGRCIRCUIT USING TRANSIMPEDANCE AMPLIFIER [7] ... 29

FIGURE 17WEAK INVERSION PMOSOPERATIONAL AMPLIFIER... 31

FIGURE 18NMOSOPERATIONAL AMPLIFIER USING LEVEL SHIFTERS... 31

FIGURE 19OPERATIONAL TRANSCONDUCTANCE AMPLIFIER... 33

FIGURE 20 THRESHOLD VOLTAGE BASED VOLTAGE REFERENCE... 34

FIGURE 21CMOSBANDGAP REFERENCE CIRCUIT... 35

FIGURE 22 PROPOSED BANDGAP REFERENCE CIRCUIT... 36

FIGURE 23VARIATION OF PN JUNCTION VOLTAGE WITH TEMPERATURE... 37

FIGURE 24CIRCUIT FOR PTAT VOLTAGE GENERATION... 38

FIGURE 25 PTAT VOLTAGE VARIATION WITH TEMPERATURE... 39

FIGURE 26BANDGAP CIRCUIT... 39

FIGURE 27 REFERENCE VOLTAGE VARIATION WITH TEMPERATURE... 41

FIGURE 28OPERATIONAL AMPLIFIER... 41

FIGURE 29 VA,VB AND VREF VARIATION WITH SUPPLY VOLTAGE... 42

FIGURE 30 A)BLOCK DIAGRAM OF SELF BIASED REFERENCE B)REGIONS OF OPERATION... 43

FIGURE 31 VREF VARIATION WITH SUPPLY VOLTAGE... 44

FIGURE 32OPERATING POINTS OF OPERATIONAL AMPLIFIER... 45

FIGURE 33STARTUP CIRCUIT... 46

FIGURE 34 A)CURRENT THROUGH TRANSISTOR M6. B)CURRENT THROUGH M4... 46

FIGURE 35BGRCIRCUIT WITH CURVATURE COMPENSATION... 48

FIGURE 36 VREF VARIATION AFTER CURVATURE COMPENSATION... 49

FIGURE 37 A)VREFVARIATION WITH TEMPERATURE B)VREFVARIATION AFTER CURVATURE COMPENSATION ... 51

FIGURE 38 VREF VARIATION WITH SUPPLY VOLTAGE... 51

List of Tables

TABLE 1CURRENT SUMMING BGR... 24

TABLE 2DTMOSBGR... 25

TABLE 3BGRBASED ON RESISTIVE SUB DIVISION... 28

TABLE 4BGRUSING TRANSIMPEDANCE AMPLIFIER... 30

TABLE 5BGRUSING DEPLETION TRANSISTORS... 30

TABLE 6THRESHOLD VOLTAGE BASED VOLTAGE REFERENCE... 34

TABLE 7OPERATIONAL AMPLIFIER FEATURES... 43

TABLE 8COMPONENT VALUES OF BGRCIRCUIT... 50

TABLE 9MEASURED RESULTS... 52

TABLE 10RESULT COMPARISON... 53

1 Introduction

Precision voltage reference circuits are necessary for accurate working of mixed and analog integrated circuits such as oscillators, PLLs, Data Converters and Dynamic Random Access Memories (DRAM’s). These voltage references should be insensitive to variations in process, temperature and supply voltage. The performance of many mixed analog/digital systems is limited by inaccuracies and power supply noise coupling errors in integrated voltage references [1, 20]. So precision voltage reference circuits forms an integral part of almost all integrated circuit designs.

Some of the desired characteristics of a voltage reference circuit are-

Silicon implementable Stable and accurate

Independent of output loading

Insensitive to power supply variations (especially for battery operated devices) Insensitive to temperature

Most popular reference voltage generators are-

Zener-based Voltage References Bandgap Voltage References

1.1 Zener Based Voltage References

Zener-based temperature compensated voltage reference circuits were popular few years back. However these devices have breakdown voltages greater than 6V, which puts a lower limit on the supply voltage requirements. Further, they require tight process control to maintain a given tolerance, and they are relatively noisy [1, 21]. So mainly because of the power supply requirements zener-based voltage references are no more popular in the latest integrated circuits.

1.2 Bandgap Voltage Reference

Bandgap voltage reference circuit uses the negative temperature coefficient of emitter base voltage in conjunction with the positive temperature coefficient of emitter-base voltage differential of two transistors operating at different current densities to make a zero temperature coefficient reference. Bandgap reference circuits gained popularity for the reasons discussed below

1.3 Advantages of Bandgap Reference circuit

The base-emitter voltage of a bipolar transistor is most predictable and well understood parameter.

Temperature insensitive.

Can operate at low supply voltages.

Capable of producing “arbitrary” output voltages.

Circuit can be easily incorporated in a monolithic IC design.

Advances in sub-micron CMOS processes resulted in supply voltages getting scaled and latest process operate at sub 1V supply voltage. Further, increased demand for battery powered portable devices and low-power designs require low supply voltages. This trend presents a great challenge in designing bandgap reference circuits which can operate at very low voltages.

Work done in this thesis is classified into two sections:-

1. Comprehensive study of existing bandgap reference circuits and understand the limitations in implementing these designs in processes with sub 1V supply.

2 Terminology and Definitions

A brief explanation of frequently used terms in this document are given here-

2.1 Bandgap Voltage

Bandgap voltage refers to the voltage difference between the valence band and conduction band of the semiconductor material, which has a constant value and its variation with temperature, is significantly less.

2.2 PTAT Voltage

PTAT stands for Proportional to Absolute Temperature Voltage, meaning the variations in voltage is proportional to temperature, or voltage increases with temperature.

2.3 CTAT Voltage

CTAT stands for Complementary to Absolute Temperature Voltage, meaning the variations in voltage is complementary to temperature, or voltage decreases with increase in temperature.

2.4 Bandgap Reference Circuit (BGR)

BGR is a precision voltage reference circuit, in which the negative temperature dependency of a voltage source is cancelled by the positive voltage dependency of another voltage source, resulting in a stable voltage at the reference temperature which is equal to the bandgap voltage of the semiconductor at the reference temperature.

2.5 V

BE

Voltage

It is the potential drop across a forward biased diode connected bipolar junction transistor (BJT)

2.6 Parts Per Million (ppm)

Reference-accuracy unit used commonly with precision voltage reference designs. Designers typically use this measure to specify temperature coefficients and other parameters that change little under varying conditions. For a 2.5V reference, 1ppm is one-millionth of 2.5V, or 2.5µV. If the reference is accurate to within 10ppm, then it is extremely good performance for any voltage reference.

3 Bandgap Voltage Reference Principle

By definition bandgap reference circuit is a voltage reference circuit, the output of which is equal to the bandgap voltage of the semiconductor used. The first bandgap reference circuit was proposed by Robert Widlar in 1971 [1]. This circuit was implemented in the conventional junction isolated bipolar technology, to make a stable low voltage reference of 1.26V, which is the bandgap voltage of silicon at room temperature (25oC).

Early implementation of these voltage references were based on the difference between the threshold voltages of enhancement and depletion mode MOS transistors [2]. This provides low temperature coefficient (TC) but the drawback being the output is not easy to control because of the direct dependence on the doses of ion implantation steps and further depletion mode transistors are not available in most of the CMOS processes.

So in modern processes the implementation principle of bandgap voltage reference is to cancel the negative temperature coefficient(TC) of a pn junction (In practice the base emitter voltage of a bipolar transistor (VBE)) with the positive temperature coefficient(TC) of a thermal voltage given by T

kT V

q

= , where ‘k’ is Boltzman’s constant, ‘T’ the absolute temperature and ‘q’ the electron charge. This principle is illustrated in Fig.1 [4].

It is well understood that the pn junction voltage (VBE) is nearly complementary to absolute temperature(CTAT) meaning it decreases (≈ -2mV/ 0 C) almost linearly with temperature [3]. It is also noted that VBE is equal to the bandgap voltage (VG0) of the semiconductor, in a first order approximation extrapolated to absolute zero. If another voltage equal in magnitude to VBE but proportional to absolute temperature (PTAT) is summed with VBE, we obtain a voltage equal to VG0. In this way a well defined voltage reference in generated which is independent of temperature. The resulting reference voltage will have a temperature coefficient of around 10ppm/ 0K.

Figure 1 Hypothetical Bandgap Reference Circuit

A PTAT voltage can be obtained through the difference of two VBEs biased at different current densities. The relation is given by

∆ VBE = VBE1 - VBE2 =VT ln C2 C1 J J ⎛ ⎞ ⎜ ⎟ ⎝ ⎠ (1)

JC2 and JC1 being the current densities of VBE junctions. Typical bipolar implementation [4] of the bandgap reference circuit is given in Fig. 2.

3.1 Derivation of the Temperature Independent Voltage

Conditions for temperature compensation can be derived starting with the V-I relationship of a forward biased base emitter junction of a bipolar transistor given by

BE qV kT c s

I

I e

⎛ ⎞ ⎜ ⎟ ⎝ ⎠

=

where IS is the saturation current of the transistor. The complete equation for the base emitter voltage (VBE) of a transistor is given by

0 0 0 0 0 1 ln G c BE BE c T T nkT T kT V V V T T q T q ⎛ ⎞ ⎛ ⎞ ⎛ ⎞ ⎛ = _⎜ − _⎟+ _{⎜ ⎟}+ _{⎜ ⎟ ⎜} ⎝ ⎠ ⎝ ⎠ ⎝ ⎠ ⎝ 0 ln I I ⎞ ⎟ ⎠ (2)

where is the extrapolated bandgap voltage of the semiconductor material at absolute zero temperature, q the charge of an electron, n is a process constant (equals 1.5 in most processes), k Boltzman’s constant, T is absolute temperature, Ic is collector current and is the base emitter voltage at temperature [4]. The last two terms in (2) can be ignored as they are quite small and can be made even smaller by making Ic vary with absolute temperature. This is the CTAT voltage.

0 G V 0 BE V T0

We have already seen how to generate a PTAT voltage and the expression for a PTAT voltage is given by ∆ VBE = VBE1 - VBE2 = VT ln C2 C1 J J ⎛ ⎞ ⎜ ⎟ ⎝ ⎠ = 2 1 ln c c kT J q J ⎛ ⎜ ⎝ ⎠ ⎞ ⎟

Now having both VBE and ∆VBE the temperature independent reference voltage is obtained by adding (1) to (2) in its simplified form, giving

2 0 0 0 0 1 l c REF G BE c T T kT V V V T T q ⎛ ⎞ ⎛ ⎞ ⎛ = _⎜ − _⎟+ _{⎜ ⎟}+ _⎜ ⎝ ⎠ ⎝ ⎠ ⎝ 1 n J J ⎞ ⎟ ⎠ (3)

Differentiating with respect to temperature yields

0 0 0 0 1 ln REF G BE c c V V V k J T T T q J ∂ _{= − + +} ⎛ ⎞ ⎜ ⎟ ∂ ⎝ ⎠ 2 (4) For zero temperature dependence, this quantity should equal zero, giving

0 0 0 1 ln c G BE c kT J V V q J ⎛ = + _⎜ ⎝ ⎠ 2⎞ ⎟ (5)

The first term on the right is the initial base emitter voltage while the second is the component proportional to the base emitter voltage difference. Hence if the sum of the two is equal to the bandgap voltage of the semiconductor, the reference will be temperature compensated. In practice, for minimum drift, it is necessary to make the output voltage somewhat higher than the theoretical value, in order to compensate for various low order terms that could not be included in the derivation. However this is only first order temperature compensation.

Now let us see how this voltage is realized using bipolar transistors (BJT) in Fig. 2. ∆ VBE (ON) = VR3 = 1 2 2 2 2 1 1 1 . . .ln .ln . . S S T T S S I I R V V I I I R ⎛ ⎞₌ ⎛ ⎜ ⎟ ⎜ ⎝ ⎠ ⎝ I ⎞ ⎟ ⎠

Then, the voltage across R2is-

2 3 2 2 3 R R V V I R = R = 2 2 2 2 3 3 1 . . .ln . S R BE T S 2 1 R R R I V V V R R R I ⎛ ⎞ = ∆ = _{⎜ ⎟} ⎝ ⎠

Giving the final voltage-

2 2 1 3 1 . . .ln . S REF BE T S 2 1 R R I V V V R R I ⎛ ⎞ = + _{⎜ ⎟} ⎝ ⎠ VREF = VBE1 + K ∆VBE (6)

(6) gives the fundamental equation of a bandgap reference circuit, where the multiplication constant K is set by the ratio of resistors R2 to R3. In the case of silicon, the value of

V

REF at 250C is 1.26V. So a bandgap reference circuit at room temperature gives

a reference voltage of 1.26V. The value of multiplication factor can be set either by resistors R2 or R3 or by the emitter area of bipolar transistors Q1 and Q2.

Intuitively, the following things can be observed from (6).

1. The temperature Coefficient (TCF) of

V

BE is negative while that of

V

T is positive.

2. Magnitude of both temperature coefficients are not equal so a multiplication factor K is introduced, which can be either set by ratio R2 of R3, or by the emitter areas of transistors Q1 and Q2.

3. Op amp ensures that the current flowing through R1 and R2 is the same. This is achieved by the feedback from output to input.

4. Eq. (6) compensates only first order temperature dependency. Second order effects arises as the variations in temperature coefficients of the terms in (6) is not linear throughout the temperature range.

5. One may wonder why we require bipolar transistors in this design, can this not be done using the MOS transistors, the reason being bipolar transistors are efficient for biasing it in very low currents in the range of µA and they also have a well defined

V

BE. If we had gone for MOS transistors then we may end up in using

very big resistors to reduce current, but thereby consuming more area.

Variation of reference voltage with respect to temperature is shown if Fig. 3.

3.2 CMOS Bandgap Reference Circuits

In most of the CMOS processes independent bipolar transistors are not available. But still most of the CMOS voltage references also make use of bandgap reference concept. For this they rely on well transistors [2]. These are vertical bipolar transistors that use wells as their bases and substrate as their collectors; these are shown in Fig. 4.

Figure 4 a) Vertical NPN transistor b) Vertical PNP transistor

These transistors have high current gains, but their main drawback is the series base resistance due to the large lateral dimensions between the base contact and the effective emitter region [5]. To minimize this error the maximum collector current is kept below 0.1mA. Another drawback is the offset voltage due to the resistors used in the design,

meaning it’s difficult to fabricate precise value resistors, which leads to large variations in the output reference voltage and thus degrading the temperature stability of the circuit [6]. Offset problem can be addressed by laser trimming, that is each resistor can be individually trimmed to the exact value during fabrication. These issues have to be addressed while designing a CMOS bandgap reference circuit. A typical CMOS bandgap reference circuit is shown in Fig. 5.

Figure 5 Typical CMOS Bandgap Reference Circuit Here the reference voltage is given by,

Vref = VBE2 + VTln 1 2 A A ⎛ ⎞ ⎜ ⎟ ⎝ ⎠ (7)

where A1 and A2 are the emitter areas of Q1 and Q2. Intuitively the first term to right in (7) is a CTAT voltage and the second term a PTAT voltage. It is also of interest to note the minimum supply voltage for the operation of this circuit [7]. VDDmin can be expressed as

VDDmin = VREF + Vdsat (PMOS) (8)

For most designs VREF = 1.26V and Vdsat (PMOS) can vary from 0.1 to 0.3V depending on the process, so theoretically the minimum supply voltage is around 1.4V.

3.3 Solving the offset error

As mentioned the main concern in CMOS bandgap references is the offset problem, this considerably reduces the accuracy of the output voltage, though laser trimming is a solution it is enormously expensive. Paper [8] shows that the dominant term of the output voltage is indeed a function of the offset voltage, so to achieve improved accuracy offset error has to be reduced. Switched capacitor operational amplifiers is a solution, another solution using a chopped operational amplifier is presented in [8]. They propose the circuit in Fig. 6.

Figure 6 Offset Error Compensation using Chopped Operational Amplifier

Transistors M10, M11, M13, and M15 act as input choppers of the applied voltage difference at IN+ and IN-. The offset from M2 and M3, the input pair, as well as the offset from the current mirror pair, M6 and M7, are cancelled by the second chopper, M19, M20, M21, and M22. Due to the transposition at the third chopper M27, M28, M29, and M30 the offset of the current sources M12 and M13 are also eliminated. Thus M31 provides the bandgap referenced output voltage. Simulations show that the use of chopping techniques reduces the spread of the output voltage to 3.2mV as compared to 32mV without chopping. The total power consumption is measured to be 7.5µW.

4 Case Study of Low Voltage CMOS Bandgap Circuits

Supply voltage is scaling and increased demand for low power portable devices has necessitated the design of sub 1V reference circuits. The BGR topologies discussed till now gives a non scalable reference voltage of approximately 1.26V. The feasibility of a sub 1V BGR is investigated in [9], where they propose two different techniques. The first technique operates by summing two currents with opposite temperature dependence on a resistor, and the resistor value further controls the reference voltage. The second technique sums two voltages that are first attenuated, where resistive voltage dividers are used for the determination of the attenuation factor. The circuit that sums two currents is given in Fig.7.

Figure 7 Current Summing BGR

Current Summing BGR is composed of three sub circuits. The first generates the PTAT current; the second mirrors the current to another transistor, which generates the CTAT component. The last sub circuit consists of a resistor whose function is to sum the currents and convert it to the desired voltage reference level. The minimum supply voltage required for correct operation is 0.7, the voltage of the forward biased diode, and the drain to source voltage of the output transistor of the current mirror driving it. The drain to source voltage can be as low as 0.2V. In this way a low voltage BGR can be realized.

Figure 8 Voltage Summing BGR

Voltage summing BGR in Fig.8 is also composed of three sub circuits. The only difference between the current summing BGR and the voltage summing BGR is the third sub circuit. The third section is composed of a differential amplifier in a non-inverting feedback loop. The offset voltage from the use of unmatched bipolar transistors generates the PTAT component. The applied diode voltage is not the full base-emitter voltage, as in a standard BGR, but a fraction. The minimum supply voltage of one path is VT plus a VCEsat, plus the source to drain voltage of the current source. The second path’s minimum supply voltage is a VBE plus the minimum voltage of the current source plus the output voltage of the VBE generator. This value is equal to 1V with the technology that was used in this study.

Results of this study are presented in table1 and variations in reference voltage with supply voltage and temperature is plotted in Fig.9. The output voltage was found to vary by less the 0.5% over the 0.9V to 2.5V range. In the same range the temperature dependence varied by 2%.

Table 1 Current Summing BGR

Technology Supply Voltage Ref. Voltage Temperature Coefficient

Figure 9 a) VREF vs. Vdd b) VREF vs. Temperature

4.1 Dynamic Threshold MOS (DTMOS) Transistors

Another technique employed in the design of low voltage BGR design is through the use of dynamic threshold MOS (DTMOS) devices [10]. As we have seen the bandgap for low power applications can be made to appear smaller through resistive subdivision, but it is at the expense of area. The bandgap voltage can also be made to appear smaller if the diode junction is in the presence of an electrostatic field. This method can be implemented by replacing the normal diodes with MOS diodes that have interconnected gates and back gates. These devices are DTMOS devices; a cross-section is shown in Fig.10. The use of a P-DTMOS device results in a VG0 of 0.6V and the temperature gradient is approximately –1mV/K. These values are half the typical values of a standard BGR. Overview of this topology is given in table 2.

Table 2 DTMOS BGR

Technology Supply Voltage Ref. Voltage Temperature Coefficient

Figure 10 DTMOS cross section

A DTMOS BGR can be designed using the same topology as that of a standard CMOS BGR. Fig. 11 explains such a circuit. The circuit consists of a folded cascode operational amplifier and matched resistors with unequal value. The DTMOS diodes are shown with the gate-back gate connection. The input stage also utilizes DTMOS transistors, which allows operation at low supply voltages. The op amp’s output stage, shaded, uses a low voltage current mirror. Correct operation of this op amp was verified for supply voltages down to 0.7V. The circuit’s temperature dependence is shown in Fig.12. The variation over the range, -20°C to 100°C, is just 4.5mV.

Figure 12 DTMOS BGR Temperature dependence

4.2 CMOS BGR Using Resistive Sub-Division

Another low voltage BGR circuit is proposed in [11]. Topology presented in this paper has been employed in many of the low voltage bandgap reference circuits with modifications. Their circuit topology is shown in Fig. 13. Here the diodes can be replaced by PnP transistors available in latest processes.

Figure 13 BGR based on resistive sub division Here the reference voltage is given by,

_ 4 2

REF R REF CONV

V V

R

where is the conventional BGR voltage. So by suitable selection of resistor values R4 and R2, lower reference voltages than the conventional one can be achieved. Experimental result for this circuit is given in Fig. 14. The circuit was designed for a reference voltage of 515mV. V

_

REF CONV V

REF showed a variation of 515mV ± 1mV for the supply variation of 2.2 to 4 V at 27°C; and 515mV ± 3mV for temperature variation from 27 to 125°C. However the minimum supply voltage was limited to 2.1V. Summary of the design is given in table 3.

Figure 14 VREF variation in [11]

Several modifications were proposed to the circuit presented in [11]. [12] proposes the use of cascode devices to improve the output impedance of the current sources, by increasing the output impedance of current sources, sensitivity of VREF to supply noise is reduced. Resistors R1 and R2 of Fig.13 are replaced with series equivalents. The addition of nodes V3 and V4 improve the ability of the op amp to operate in sub 1V conditions. The proposed circuit is given in Fig. 15. The circuit was simulated through 20°C to 100°C and supply voltages of 0.95V to 1.50V. The curves are found to have a spread of less than 0.24%.

Table 3 BGR Based on Resistive sub division

Technology Supply Voltage Ref. Voltage Temperature Coefficient

Figure 15 BGR Circuit with improved noise Immunity

4.3 BGR Using Transimpedance Amplifier

[7] further modified the circuit in [11] by using resistors in place of the input differential stage of the operational amplifier. This circuit is given in Fig. 16.

Figure 16 BGR Circuit Using Transimpedance Amplifier [7]

Here they obtain a PTAT current by sensing the voltage difference; this current is summed with a current complementary to VBE to obtain the reference voltage. This is achieved with the help of a transimpedance amplifier. Vref of this circuit is given by

1 2 3 1 2 2 1 . . ln BE b REF A V V V R VT 2 R A R R ⎡ ⎛ ⎞ ⎤ = _{⎢ ⎜ ⎟}+ _⎥ ⎝ ⎠ ⎣ − ⎦

Similar to [11] the value of Vref can be changed by choosing different values of R1, R2, and R3. Experimental results for a Vref of 1V were shown to be accurate within ±1% over 0°C to 100°C, with R1 untrimmed and ±0.3% after trimmed. Summary of the design given in table 4.

Table 4 BGR Using Transimpedance Amplifier

Technology Supply Voltage Ref. Voltage Temperature Coefficient

1.2µm BiCMOS 1.4V 1.2V 7ppm

4.4 BGR Using Depletion Transistors

[13] also suggests improvements to the circuit in [11]. The motivation for the improvements is that the differential amplifier in [11] has MOS depletion transistors in the input stage. These devices are not used in standard processes and result in higher process costs. PMOS transistors in weak inversion are proposed in place of the depletion mode transistors of [11]. The new circuit is shown in Fig. 17. As the supply voltage is decreased below 1.4V the input devices enter weak inversion. The BGR remains biased correctly as long as the supply is above 0.9V, below which the loop gain is insufficient and the operational amplifier fails to function.

Table 5 BGR Using Depletion Transistors

Technology Supply Voltage Ref. Voltage Temperature Coefficient

Figure 17 Weak Inversion PMOS Operational Amplifier

A second circuit (Fig. 18) proposed uses an NMOS topology with PMOS level shifters to provide the correct common mode voltage to the input stage. The circuit is unable to operate with a supply voltage as low as the first, but it does not require a startup circuit for correct biasing at power-on. The supply voltage is limited to approximately 1.4V. Summary of the design given in table 5.

Figure 18 NMOS Operational Amplifier using Level Shifters

Studying these low voltage bandgap reference circuits following things can be observed. The technique used for the design of conventional bandgap circuits can also be extended to the design of low voltage temperature independent circuits. Still the circuits bear the

name bandgap, though the reference voltage is not equal to the silicon bandgap because the idea used for implementing these low voltage circuits is same that of conventional design, i.e. to cancel the CTAT dependency with a PTAT dependency. Sub 1V BGR designs require a minimum supply voltage of 700mV, which is the saturation voltage of the diode connected transistors used (BJT). Further this minimum voltage is also dependent on the operational amplifier design. We need to ensure that this minimum voltage is sufficient for the proper operation of the operational amplifier. Careful understanding of the circuits reveals that low voltage reference circuit designs are mainly limited by the voltage requirements of the operational amplifier. Paper [12] is an example. We have already seen, this topology can give very low temperature independent reference voltage, but the operational amplifier requires a minimum supply voltage of 1.5V. So design of operational amplifiers which can give significant gain at low supply voltages should be the main design motto.

Design of low voltage reference circuits has start up issues. Since the voltage levels are low we need to ensure that the circuit is getting properly biased on start up. This start up circuits should be designed in such a way that once the circuit gets stabilized, it should not further influence the circuit operation.

Circuits discussed till now compensates first order temperature dependency, second order dependency arises as the variation in the junction voltage is not linear through out the operating temperature range. So to compensate second order effects additional circuitry has to be incorporated in these designs thereby giving high precision voltage references. So to conclude this study, the main disadvantages of BGR circuit are given below.

4.5 Disadvantages of BGR Circuit

Additional blocks needed to ensure a non-zero output voltage viz. star-up Accuracy of models is limited- trial and error technique required

4.6 BGR Circuit Using Bulk Biasing

If we want designs to operate below 700mV, then with existing processes it is not possible as the BJT requires at least 700mV to remain in saturation. Further as we don’t have well defined voltages, designs based on conventional techniques are not possible. Alternate techniques have to be explored. Paper [14], [15] discusses issues related to design of voltage references that can work at sub 0.6V supply.

Paper [14] discusses bulk biasing to reduce the threshold voltage of PMOS and NMOS transistors and it uses operational transconductance amplifier (OTA) in place of the conventional operational amplifier. Fig. 19 shows the proposed OTA. In this circuit, vertical bipolar transistors are replaced by MOSFETs operating in the weak inversion mode, thereby reducing the VBE voltage drop from 0.7 to 0.2. Bulk biasing is the technique by which we forward bias the source bulk junction to reduce the threshold voltage.

4.7 Threshold Voltage Based Voltage Reference

Paper [15] introduces the new concept of threshold voltage based voltage references. The basic idea here is to compensate the temperature dependency of the threshold voltage (VT) of a PMOS transistor with that of an NMOS transistor, both having a CTAT dependency. This concept is pictorially shown in Fig. 20. The performance of the proposed voltage reference is shown to be comparable to bandgap circuits, but at the cost of more area and complexity.

Figure 20 Threshold Voltage Based Voltage Reference

Table 6 Threshold Voltage Based voltage Reference

Technology Supply Voltage Ref. Voltage Temperature Coefficient

5 Proposed Low Voltage Low Power BGR

Proposed design is motivated by the requirement of a precision voltage reference that can work at low supply voltage. It also incorporates a simple technique to compensate for the second order temperature dependency. The circuit topology is quite similar to the one discussed in [18], which was implemented in a 0.8µm BiCMOS process. The design is optimized for low power, low voltage applications.

Let us now consider a typical CMOS bandgap reference circuit as given in Fig. 21.

Figure 21 CMOS Bandgap Reference Circuit

Its working is explained in the briefest terms. This circuit relies on two bipolar transistors (Q1 and Q2) working at different emitter current densities. The rich transistor (Q2) will run at typically 8 times the current density of the lean one (Q1). This factor of 8 will cause typically around 60mV delta (∆VBE) between the base emitter voltages of these two transistors. This delta voltage is amplified by a factor of around 5 and added to the forward biased pn junction voltage (VBE). These two voltages add up to 1.25V, which is approximately the bandgap voltage of silicon at room temperature given by

Vref = VBE2 + VTln 1 2 A A ⎛ ⎞ ⎜ ⎟ ⎝ ⎠ (9)

where A1 and A2 are current densities of transistors Q1 and Q2. As we have already seen the first term to the right in (9) is a CTAT voltage while the second term is a PTAT voltage.

5.1 Bandgap Reference Circuit

Fig.22 shows the proposed bandgap reference circuit. It is different from the conventional design (Fig.21.), as the PTAT voltage generated across R1 is mirrored to a diode connected transistor B4 to generate a temperature independent reference voltage, whose value can be easily set by the resistor ratio of R3 to R2.

Figure 22 Proposed Bandgap Reference Circuit

5.2 Working

Let us break the circuit and see the working part by part. The variation in voltage with respect to temperature across a pn (VBE) junction is given by

ln 2 T C T g S S BE T V V I V E V T T I I _kT ∂ ∂ = − − ∂ ∂

where VT is the thermal voltage given by VT = kT / q, where k is Boltzman’s constant, T the absolute temperature and q the electron charge [5]. In this design for a supply voltage of 1V and temperature at 25 oC VBE = 700mV giving

BE V T ∂ ∂ ≈ -1.5mV⁄ o_K.

This variation is shown in Fig. 23. This negative TC voltage has to be compensated by a positive TC voltage.

Figure 23 Variation of pn junction voltage with temperature

5.2.1 PTAT Voltage Generation

This positive TC voltage comes from the voltage difference of two pn junctions (VBE1 and VBE2) having different current densities, given by

1 2

ln

C

ln

C

ln

BE BE BE T T T S S

I

I

V

V

V

V

V

V

I

nI

−

= ∆

=

−

=

n

, (10) where n is the current density ratio of B2 to B1 [5]. In our design this is achieved through an emitter area ratio of 8 and setting the currents through B1 and B2 equal. Operational amplifier sets the emitter currents of B1 and B2 equal. Therefore voltage across R1 equals∆VBE and the current flowing through R1 is proportional to temperature (PTAT). This set up is shown in Fig. 24.

Figure 24 Circuit for PTAT voltage Generation

To reduce power, the current flowing through transistor (B1 and B2) is limited to 1µA. The high gain of the operational amplifier sets the voltage across nodes A and B equal giving,

VA = VB = VBE1 = VBE2 + I1 x R1 where VBE1 = VTln Ic

Is

⎛ ⎞

⎜ ⎟

⎝ ⎠

Therefore VBE1- VBE2 = VTln (n) = I1 x R1. Where n is 8. So the voltage across R1 is the positive TC voltage. The current across R1 is

1 ln

T

V n

R which is proportional to absolute

temperature (PTAT) current. So the current can be easily controlled by adjusting the value of R1. Variation of PTAT voltage with temperature is shown in Fig.25. It’s interesting to note that the variation with temperature in PTAT voltage is not at the same rate as that of the CTAT voltage.

Figure 25 PTAT voltage variation with Temperature

The complete circuit to generate bandgap reference voltage is shown in Fig. 26.

So the PTAT current flowing through resistor R1 is mirrored to diode connected transistor B4 using current mirror M4. The output voltage is formed across R2 and B4 by adding the positive TC voltage (I2R2) to the negative TC voltage (VBE4), resulting in a temperature independent voltage at the reference temperature of 25oC. So the reference voltage (voltage across R2 and B4) is given by

4 2 1 ln( ) 1.26 T REF BE V n V V R V R = + = (11)

using (10), equation (11) can be rewritten as

VREF = VBE4 + K∆VBE (12)

where K is set by the ratio of R2 to R1. Further VREF can also we written as

VREF = VBE4 + I2R2 (13)

The base emitter voltage is 0.7V at 25oC. We can set the PTAT current flowing through R2 to be 1µA. Therefore, from equation (13) the resistance R2 can be determined by

2 1.26 0.7 560 0.1

R = − ≈ K

This is the first order compensation. The complete equation for the reference voltage is given by [5]

(

) (

)

0 0 1 0 0

1

l

REF G

T

BE G

kT

T

V

V

V

V

m

KV

T

η

q

=

+

−

+

+ −

+

n n

₍₁₄₎

where VG0 is the bandgap voltage of silicon at 0o K, m is the temperature constant ≈ 2.3

and ηis the correction term due to temperature dependency of the diffusion resistors used, T0 is the reference temperature and VBE1 is the junction voltage of B1 at reference

temperature. The addition of resistor R3 further changes the reference voltage value to

(

4 4

)

'

REF BE M

V

=

K V

+

I

R

2 (15) where 3 2 3 ' R K R R =

+ . So by setting the value of R3 the reference voltage can be easily set.

For this design the reference voltage is set as 0.730V. Reference voltage variation with temperature is shown in Fig. 27.

Figure 27 Reference Voltage Variation with Temperature

5.2.2 Operational Amplifier

We have already seen design of low voltage bandgap voltage reference circuits are constrained by the power supply requirements of the operational amplifier. In this design operational amplifier is designed to operate at sub 1V. Fig.28. shows the circuit schematic of the operational amplifier.

It is composed of transistors M8 to M14. The purpose of the operational amplifier is to force node A and B in Fig.26 to same voltage. This is achieved by the high gain of the circuit and the feed back mechanisms that is formed around it. The operational amplifier has nmos input stage as the input voltage is nearer to Vdd, with current mirror load. The output Vb provides the bias for the entire circuit and a feedback loop is also formed. This ideally provides constant Vgs to all pmos transistors and a constant current can be obtained. The transistor channel lengths are kept fairly large to reduce the effect of noise on the output voltage. This design requires a minimum supply voltage of 900mV as shown in Fig. 29. Operational amplifier details are given in table 6.

Figure 29 Va, Vb and VREF variation with Supply Voltage

In Fig. 29 voltages at node A (Va), B (Vb) and reference voltage (VREF) is plotted against supply voltage. It is evident from the figure that these voltages take a steady level at a supply voltage of 900mV. Intuitively only at 900mV, VA equals VB.

Table 7 Operational Amplifier Features

Parameter Value

DC Gain 50dB

Gain Bandwidth Product 1Mhz Supply Voltage 900mV Offset voltage 500µV

If supply rejection was the design issue we could have added a cascode stage to the operational amplifier design, but this would have resulted in higher supply voltage requirement in order to maintain transistors in saturation.

5.2.3 Power Supply Independent Biasing

Another requirement from a reference voltage generator is the power supply independence. By which we mean, the reference voltage should not vary with changes in supply voltage. In our design power supply sensitivity is reduced by so called bootstrap bias technique also called self biasing [4]. Here instead of developing the input current by connecting a resistor to the supply, the input current is made to depend directly on the output current of the current source. This concept is illustrated in Fig. 30.

Here the important variables are the input current Iin and out put current Iout. From the standpoint of the current source, the output current is almost independent of the input current for a wide range of input currents as shown in Fig. 30b. For current mirror Iin is made equal to Iout as the gain of current mirror is set to unity. So from Fig. 30b it is evident that the circuit has two operating points A and B. A is the desired operating point and B is the undesired one as Iin = Iout = 0.

If the output current in Fig. 30a increases for any reason, the current mirror increases the input current by the same amount because the gain of the current mirror is assumed to be unity. As a result, the current source increases the output current by an amount that depends on the gain of the current source. Therefore, the loop responds to an initial change in the output current by further changing the output current in a direction that reinforces the initial change. In our design, the current source discussed here is same as the output of the operational amplifier and the current mirrors being the pmos transistors M1, M2, M3 and M4 in Fig. 22.

In practice, point B in Fig.30b is a stable operating point. Thus, unless precautions are taken, the circuit may operate in the zero current condition even when the power supply voltage is non-zero. This is a drawback of the self-biased circuit. So a startup circuit is usually required to prevent the self-biased circuit from remaining in the zero current state. Dependency of reference voltage on supply voltage is shown in Fig. 31.

5.2.4 Startup Circuit

In order to ensure proper working of the circuit, we need a mechanism which can provide a small current to flow through the operational amplifier and enable it during start up. This is required as we are operating the circuit at low supply voltage [4]. The V-I characteristic of the operational amplifier is shown in Fig.32. It can be seen that the circuit has two stable operating points. One in which no current is flowing through the circuit and the second after the transistors are saturated. So it is possible that during start up, the circuit may get biased in the operating point, where no current flows through the circuit. But this can be prevented if we can make a small current to flow through the circuit during start up.

Figure 32 Operating Points of Operational Amplifier

This is achieved through transistors M5 to M7 in Fig. 33. M5 is a diode connected pmos transistor which is in saturation; it provides sufficient gate voltage for M6 to turn on. When M6 is on, a small current flow through the operational amplifier because transistors M5 and M6 pull down the output node of the operational amplifier, enabling the entire circuit. Once current starts flowing, M7 is turned on and sinks all current from M5 and disables M6.

Figure 33 Startup Circuit

Transistors M5, M6, and M7 are made weak, so that it won’t influence the operation of the circuit once the BGR circuit is properly biased. This is shown in Fig. 34.

From Fig.34 it is evident that transistor M4, pulls down the output of the operational amplifier during startup, thereby making a small current to flow through the operational amplifier and once the circuit is properly biased as shown by Fig.34b it can be noticed that the current flow through transistor M4 is negligible. This model of startup circuit has the drawback as it consumes static power, but is quite simple to implement.

5.2.5 Curvature Compensation

Eq. (12) compensates temperature dependence of the output voltage at the first order only. Second order effects arises as the variation in VBE is not linear with temperature throughout the operating range, but it varies according to the relation proposed in [14] and is given by

( )

(

0

)

(

)

0 ln 0 BE G G BE T T V T V V V V T η α T = − − − − T (16)

where η is a process parameter and is 4 for this standard CMOS process, while αequals 1 if the current in BJT is PTAT and 0 when the current is temperature independent. Bandgap architecture shown in Fig.26 corrects the first term in (16) leading to second order temperature dependence. These effects are generally compensated using operational amplifiers or switched capacitor structures. However for our design we have considered techniques given in [10] and [11], as it occupies less area and easy to implement. This implementation requires an additional current mirror M3, resistors R4, R5 and BJT B4 as shown in Fig. 35.

Figure 35 BGR Circuit with Curvature Compensation

Here the idea is to correct the non linear term by a proper combination of the VBE across a junction with temperature independent current (α=0) and the VBE across a junction with a PTAT current (α=1). From Fig.35 it is evident that current in bipolar transistors B1 and B2 is PTAT (α=1), while the current in pmos transistor M3 is first order temperature independent. If we can inject this current to a diode connected bipolar transistor B3, across B3 we produce a VBE where α≈ 0 [11]. So using (16) VBE across B3 and B1, 2 can be expressed as

, 3 0 0 0 ( ) ( ) ln BE B G G BE T T T V T V V V V T η T = − − − (17) and , 1, 2 0 0 0 ( ) ( ) ( 1) ln BE B G G BE T T V T V V V V T η T = − − − − T (18) the difference of (17) and (18) lead to the voltage proportional to the non linear term of

(16) given by 3 1, 2 0 ( ) ( ) ln NL BE BE T T V V T V T V T ≅ − = (19)

where, is the voltage proportional to the non-linear term of (16). Now to achieve curvature compensation current proportional to V

NL V

NL is subtracted from the reference voltage using resistors R4 and R5, which drain current proportional to VNL, leading to the new equation.

(

4 4 2

)

' REF BE M V =K V +I R + 4, 5 1 NL R V R . (20) The value of R4, R5 which leads to curvature compensation is given by [11]

1 4, 5 1 R R η = − .

The variation of reference voltage with temperature after curvature compensation is shown in Fig.36.

Figure 36 VREF variation after curvature compensation

5.3 Design Choices and Simulated Results

Components B1 to B4 are diode connected vertical PNP transistors readily available in standard CMOS process. Resistors R1 to R5 are implemented using n well diffusion resistors. The channel lengths of PMOS transistors are kept large to reduce the effect of noise and power supply variations on reference voltage. Transistor M4 is designed twice as large as M2 and M1, thus reducing the size of resistors R2 and R3, because of more

current drive. The values of components are presented in table7. The values are optimized using Cadence® Affirma™ Simulation Tool.

Table 8 Component Values of BGR Circuit

Component Name W/L Value in µ meter

M1,M2,M3 240/7 M4 500/7 M5,M6 2/8 M7 2/5 M8,M9 25/7 M10,M11 20/4 M12,M13 25/4 M14 50/7 R1 88K R2 516K R3 900K R4 25K R5 35K Q1,Q2,Q3,Q4 (Emitter Area) 1,8,1,8

The variation in reference voltage of the BGR with and without curvature compensation is given in Fig.37. The temperature variation is measured for a 900mV supply and for a temperature range of 0 to 60 oC. The circuit achieves a temperature coefficient of 7ppm/oK. The variation in Vref with supply voltage is plotted in Fig. 38. The supply voltage is varied from 0.9 to 1.5V. The supply voltage dependence is found to be 300ppm/V. This can be further improved if we can incorporate a cascode stage in the operation amplifier design, but at the cost of a higher supply voltage, which was not the focus of this design.

Figure 37 a)VREF Variation with temperature b)VREF Variation after curvature compensation

Figure 38 VREF variation with supply voltage

Variation in reference voltage with temperature, for supply voltage in the range 0.9-1.5V is plotted in Fig 39. Summary of the measured results are given in table 8. Further improvement in performance is possible if we increase the transistor sizes and reduce the offset error of the operational amplifier.

Figure 39 Reference voltage variations for different supply voltages. Table 9 Measured Results

Parameter Measured Value

Supply voltage range 0.9-1.5V

Vref 0.730V Power consumption 5µW Temperature Variation(0-60oC) Without compensation With Compensation 400µV →14ppm/K 200µV →7ppm/K Dependence on Supply voltage 220µV/V→300ppm/V

5.4 Conclusion

In this work effort is made to understand various topologies of the existing bandgap reference circuits and identify the limitations why these circuits are difficult to be used with current processes, mainly with sub 1V technologies. Through the ideas obtained out of this study, we have designed and analyzed a curvature compensated fully CMOS bandgap reference circuit, in 0.18µm standard CMOS process. The simulated results are presented. Results show substantial reduction in power when compared to the reference design, which was implemented in 0.8µ meter BiCMOS technology. The circuit achieves a temperature coefficient of 7ppm//oK and supply voltage dependence of 300ppm/V. Comparison of the results with the reference design is presented in table 9.

Table 10 Result Comparison

Parameter Measured Reference Design Current Value

Power Supply Voltage 1.5V 900mV

Technology 0.8µm BiCMOS 0.18 µm CMOS

Power consumption 92µW 5µW

Temperature Variation(0-60oC) Without compensation

With Compensation 800µV →20ppm/K 300µV →7.5ppm/K 400µV →14ppm/K 200µV →7ppm/K Dependence on Supply voltage 114µV →212ppm/V 220µV →300ppm/V

6 References

1. Pease, Robert, The design of Bandgap Reference Circuits- Trials and Tribulations, IEEE 1990 Bipolar Circuits and Technology Meeting, p 214-218, 1990.

2. Nicollini, Germano and Senderowicz, Daniel, A CMOS Bandgap Reference for Differential Signal Processing, IEEE JSSC, vol. 26, pp 41, January 1991.

3. Lee, Thomas, The Design of CMOS Radio-Frequency Integrated Circuits, Cambridge University Press, p 237- 235, 1998.

4. Gray, Hurst, Lewis, Meyer, Analysis and Design of Analog Integrated Circuits- Fourth Edition, p 317-325 , John Wiley publication.

5. David A. Johns and Ken Martin, Analog Integrated Circuit Design, John Wiley and Sons, p 354 – 364, 1997.

6. Ferro, Marco et.al., A Floating CMOS Bandgap Voltage Reference for Differential Applications, IEEE JSSC, vol. -24, p 690-691, June 1989.

7. Jiang, Yueming, Lee, Edward, Design of Low Voltage Bandgap Reference using Transimpedance Amplifier, IEEE TCAS II, vol. 47, p552-555, June 2000.

8. Sanduleanu, M.A.T. et al., Accurate Low Power Bandgap Voltage Reference in 0.5µm CMOS Technology, Electronic Letters vol.34, p 1025-1026, May 1998. 9. Ripamonti G. et al., Low Power – Low Voltage Bandgap References for Flash

EEPROM Integrated Circuits, Design Alternatives and Experiments, Proceedings of ICES 1999, vol.2, p 635-638, 1999.

10. Annema, Anne- Johan, Low Power Bandgap References Featuring DTMOST’s, IEEE JSSC, vol.34, p 670-674, July 1999.

11. H. Banba, Hi. Shiga, A.Umezawa, T. Miyaba, T. Tanzawa, et al., A CMOS Bandgap Reference Circuit with Sub 1V Operation, IEEE Journal of Solid State Circuits, vol. 34, p 670-674, May 1999.

12. Waltari, Mikko, and Halonen, Kari, Reference Voltage Driver for Low-Voltage CMOS A/D Converters, Proceedings of ICECS 2000, vol. 1, p 28-31, 2000. 13. Pierazzi, Andrea, et al., Bandgap Reference for near 1V operation in standard

CMOS Technology, IEEE2001- Custom Integrated Circuits Conference, p 463-466, 2001.

14. Y. Dai, D.T. Comer, D.J. Comer, C.S. Petrie, Threshold Voltage Based CMOS Voltage Reference, IEE Proc.-Circuits Devices Syst., Vol. 151, No. 1, February 2004.

15. T. Yetterdal, CMOS Bandgap Voltage Reference Circuits for Supply Voltages Down to 0.6V, Electronic letters, Vol. 39, No. 20, October 2003.

16. Y.Tsividis, Accurate analyses of temperature effects in IC- VBC characteristics with application to bandgap reference sources, IEEE J. Solid State Circuits, vol. 15, p 1076-1084, Dec 1980.

17. M. Gunawan, G. meijer, J. Fonderie and H. Huijsing, A Curvature corrected Low voltage bandgap reference, IEEE J. Solid State Circuits, vol 28, p 667- 670, June 1993.

18. Piero Malcovati, Franco Maloberti, Carlo Fiocchi and Marcello Pruzzi, Curvature Compensated BiCMOS Bandgap with 1V Supply Voltage, IEEE J. Solid State Circuits, vol.36, p 1076 -1081, July 2001.

19. K.Lasanen, V. Korkala, E. Raisanen-Ruotsalainen, J. Kostamovaara, Design of a Low Power 1V CMOS Bandgap Reference Based on Resistive Subdivision, IEEE Circuits and Systems, vol. 4, p 564-567, May 2002.

20. Todd L. Brooks, Alan L Westwick, A Low-Power Differential CMOS Bandgap Reference, IEEE-ISSCC-1994.

21. Robert J. Widlar, New Developments in IC Voltage Regulators, IEEE-ISSCC, vol-6, 1971.

Copyright

Svenska

Detta dokument hålls tillgängligt på Internet – eller dess framtida ersättare –under 25 år från publiceringsdatum under förutsättning att inga extraordinära omständigheter uppstår. Tillgång till dokumentet innebär tillstånd för var och en att läsa, ladda ner, skriva ut enstaka kopior för enskilt bruk och att använda det oförändrat för ickekommersiell forskning och för undervisning. Överföring av upphovsrätten vid en senare tidpunkt kan inte upphäva detta tillstånd. All annan användning av dokumentet kräver upphovsmannens medgivande. För att garantera äktheten, säkerheten och tillgängligheten finns lösningar av teknisk och administrativ art.

Upphovsmannens ideella rätt innefattar rätt att bli nämnd som upphovsman I den omfattning som god sed kräver vid användning av dokumentet på ovan beskrivna sätt samt skydd mot att dokumentet ändras eller presenteras i sådan form eller i sådant sammanhang som är kränkande för upphovsmannens litterära eller konstnärliga anseende eller egenart.

För ytterligare information om Linköping University Electronic Press se förlagets hemsida http://www.ep.liu.se/

English

The publishers will keep this document online on the Internet – or its possible replacement – for a period of 25 years from the date of publication barring exceptional circumstances.

The online availability of the document implies a permanent permission for anyone to read, to download, to print out single copies for your own use and to use it unchanged for any non-commercial research and educational purpose. Subsequent transfers of copyright cannot revoke this permission. All other uses of the document are conditional on the consent of the copyright owner. The publisher has taken technical and administrative measures to assure authenticity, security and accessibility.

According to intellectual property law the author has the right to be mentioned when his/her work is accessed as described above and to be protected against infringement.

For additional information about the Linköping University Electronic Press and its procedures for publication and for assurance of document integrity, please refer to its www home page: http://www.ep.liu.se

© Kevin Tom 1st November, 2004.