High Level Ultra Low Power Transmitters for the MICS Standard

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Institutionen för systemteknik

Department of Electrical Engineering

Examensarbete

A Study on QPSK Modulator Architectures for Ultra

Low Power Transmitters

Examensarbete utfört i Elektroniska komponenter vid Tekniska högskolan i Linköping

av

Per Eidenvall, Nils Gran

LiTH-ISY-EX--10/4357--SE

Linköping 2010

Department of Electrical Engineering Linköpings tekniska högskola

Linköpings universitet Linköpings universitet

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A Study on QPSK Modulator Architectures for Ultra

Low Power Transmitters

Examensarbete utfört i Elektroniska komponenter

vid Tekniska högskolan i Linköping

av

Per Eidenvall, Nils Gran

LiTH-ISY-EX--10/4357--SE

Handledare: Atila Alvandpour

, Linköpings universitet

Examinator: Atila Alvandpour

, Linköpings universitet

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Avdelning, Institution

Division, Department Division of Electronic Devices Department of Electrical Engineering Linköpings universitet

SE-581 83 Linköping, Sweden

Datum Date 2010-11-22 Språk Language Svenska/Swedish Engelska/English Rapporttyp Report category Licentiatavhandling Examensarbete C-uppsats D-uppsats Övrig rapport

URL för elektronisk version

http://www.ek.isy.liu.se http://urn.kb.se/resolve?urn=urn:nbn:se:liu:diva-62656 ISBN — ISRN LiTH-ISY-EX--10/4357--SE

Serietitel och serienummer

Title of series, numbering

ISSN

—

Titel

Title A Study on QPSK Modulator Architectures for Ultra Low Power Transmitters

Författare

Author

Per Eidenvall, Nils Gran

Sammanfattning

Abstract

Today, medical implants such as cardiac pacemakers, neurostimulators, hearing aids and drug delivery systems are increasingly more important and frequently used in the health care system. This type of devices have historically used inductive coupling as communication medium. New demands on accessibility and increased performance in technology drives new research toward using radio communications. The FCC MICS radio standard are specifically devoted for implantable devices.

Basically all published research on transmitters in this area are using frequency shift keying (FSK) modulation. The purpose of this thesis is to explore the viability of using phase shift keying (PSK) modulation in ultra low power transmitters and suggest suitable architectures.

Nyckelord

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Abstract

Today, medical implants such as cardiac pacemakers, neurostimulators, hearing aids and drug delivery systems are increasingly more important and frequently used in the health care system. This type of devices have historically used in-ductive coupling as communication medium. New demands on accessibility and increased performance in technology drives new research toward using ra-dio communications. The FCC MICS rara-dio standard are specifically devoted for implantable devices.

Basically all published research on transmitters in this area are using frequency shift keying (FSK) modulation. The purpose of this thesis is to explore the viability of using phase shift keying (PSK) modulation in ultra low power transmitters and suggest suitable architectures.

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Acknowledgments

We would like to thank:

Atila Alvandpour, Professor

Håkan Bengtsson at Zarlink Semiconductor Inc. Amin Ojani, Ph.D. student

Jonas Fritzin, Ph.D. student Johan Bengtsson

for their help in this thesis

Per Eidenvall and Nils Gran

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diagram for a data rate of 250 kbit/s. 4 intermediate steps are shown in (a) and (b) and 8 are shown in (c) and (d). . . 53 7.10 Ramp-up ratio for the direct multiplexing architectures. . . 53 7.11 Frequency spectrum for the direct multiplexing architectures. . . . 54 8.1 Graph showing what data rates could be considered for which

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xvi Abbreviations and Acronyms

Abbreviations and Acronyms

Acronyms Explanation

ADS Advanced Design System (Agilent)

AWGN Additive White Gaussian Noise

bps Bit Per Second

CB Radio Citizens Band Radio

Coherent detection Phase aligned- or synchronized detection

CPFSK Continuous Phase Frequency Shift Keying

CPM Continuous Phase Modulation

DCO Digital Controlled Oscillator

DQPSK Differential Quadrature Phase Shift Keying

EIRP Equivalent Isotropically Radiated Power

FLL Frequency Locked Loop

FRS Family Radio Service

FSK Frequency Shift Keying

FSM Finite State Machine

GMSK Gaussian Minimum Shift Keying

I In-phase Signal

LO Local Oscillator

LPRS Low Power Radio Service

MICS Medical Implantable Communications System

MSK Minimum Shift Keying

MURS Multi-Use Radio Service

OQPSK Offset Quadrature Phase Shift Keying

PA Power Amplifier

PFD Phase Frequency Detector

PLL Phase Locked Loop

PSK Phase Shift Keying

Q Quadrature Signal

QAM Quadrature Amplitude Modulation

QPSK Quadrature Phase Shift Keying

R/C Radio Control (also Radio Control Radio Service)

RBW Resolution Bandwidth

RF Radio Frequency

ROM Read Only Memory

SNR Signal to Noise Ratio

VCO Voltage Controlled Oscillator

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Chapter 1

Introduction

This chapter presents the introduction of the thesis. The introduction describes the purpose, background, goal, method and delimitations and gives a report outline.

1.1 Purpose

The division of Electronic Devices at the Department of Electrical Engineering at the Linköping University has a long history of researching and working with high-speed and low power CMOS devices. Recently the division has turned focus toward ultra-low power radio transmitters used in e.g. medical implants. Today basically all published research on transmitters in this area are using frequency shift keying (FSK) modulation. The purpose of this thesis is to explore the viability of using phase shift keying (PSK) modulation in ultra-low power transmitters and suggest suitable architectures.

1.2 Background

Today medical implants such as cardiac pacemakers, neurostimulators, hearing aids and drug delivery systems are increasingly more important and frequently used in the health care system. The implant must be able to communicate with the outside world to enable evaluation and configuration of its performance and to study medical events that a patient is experiencing or has experienced. Tra-ditionally communication, i.e. transmission of information, was carried out by magnetic coupling1_{[19]. With magnetic coupling a reader/transmitter head must}

in general be placed directly on the patients skin in the absolute proximity of the implant. The reading of data from the device takes relative long time. The data rate could range about 50 kbit/s [19] over a distance of only a few inches. The benefit of using magnetic coupling is the relatively low power consumption.

1_{Also often referred to inductive coupling.}

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2 Introduction

Today new demands on accessibility and advances in technology drive the attention toward radio transmission instead of magnetic coupling. Radio has the benefit of higher data rates and longer transmission distances. Radio is not affected by electromagnetic interference which inductive coupling is. Thereby the safety of the patient is improved. The major challenge is however to achieve the same low power consumption as the magnetic coupling. Because of the difficulty of changing or recharging the power source of medical implants, the power consumption is a major issue.

1.3 Goal

The goals of this thesis are:

1. Provide a summary on current ultra-low power transmitters for the MICS standard in terms of modulation scheme, power consumption, data rate and output power.

2. Suggest high level, mixer-less, PSK architectures relevant for future work. 3. Describe the benefits and drawbacks of using PSK in ultra-low power

trans-mitters used in compliance with the MICS standard.

1.4 Method

In order to achieve the goals presented in the previous section 1.3, a method has been designed. The method is divided into the following three phases.

Phase 1: Perform a literature study on current ultra-low power designs to gather specifications and give inspiration for new ideas to be used in the next phase. The study should be based on the most relevant IEEE papers.

Phase 2: Design and implement mixer-less high level phase shift keying architec-tures for computer simulations in ADS.

Phase 3: Simulate and evaluate the implemented PSK architectures.

1.5 Delimitations

1.5.1 High Level Architectures

This thesis is restricted to the investigation and design of high level phase-shift-keying architectures for simulation in ADS viable for ultra-low power operation. Hence, this thesis does not give any suggestions or examples of on chip imple-mentations. This is left for future work.

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1.6 Tools 3

1.5.2 Mixer-Less Architectures

At the beginning of the work a decision was made to focus on mixer-less architec-tures. Mixers typically require linear signal paths both at their inputs and outputs when used in PSK transmitters with phase shaping. This also affects the power consumption of the power amplifier since linear PA:s tends to consume more power than their non-linear counterpart. Mixers were also recognized as a main contributor to the global power consumption. For these reasons it was decided to narrow the field and focus on mixer-less architectures.

1.5.3 MICS Standard

The MICS standard has become the major radio standard for medical implants since FCC introduced it in 1999 (see chapter 4). Most current papers on ultra-low power radio transmitters for medical implants states that they conform to the MICS standard to some degree. This thesis will for this reason focus primarily on PSK transmitters complying with the MICS standard.

1.6 Tools

The following tools has been used during the work:

ADS 2009 Advanced System Design from Agilent has been the main tool during the work. All architectural models have been designed and simulated in ADS. Ptolemy functional blocks, i.e. blocks from the DSP design type, have been used when possible in order to minimize simulation time.

Matlab 7.7.0 (R2008b) Matlab has been used for verifying our models used in ADS and calculating appropriate input parameters. Matlab has also been used to generate many of the plots in the thesis. Appart from standard functions, functions from the System Toolbox have been used.

1.7 Report Outline

Chapter 1 presents the introduction of the thesis. The introduction describes the purpose, background, goal, method and delimitations and gives a report outline.

Chapter 2 cover some of the basic theory related to this thesis. The chapter discuss: common trade-off issues in radio frequency (RF) design related to bandwidth, power and data rate; frequency synthesis using charge pump based phase locked loops; minimum tone spacing in FSK systems; spectral regrowth issues and constant envelope behavior. The chapter is foremost intended for the novice RF reader.

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4 Introduction Chapter 3 presents an overview of ten low-power transmitters, a description of

the three most power-efficient architectures and a short discussion on using 65 nm technology in MICS applications.

Chapter 4 presents a summary of the Medical Implantable Communications Ser-vice (MICS) radio standard introduced by Federal Communications Com-mission (FCC). It also points out bandwidth efficiency issues for PSK mod-ulation complying to the MICS standard, which will be a major challenge throughout the thesis.

Chapter 5 describes the two testbenches used during simulations, the available measurements and the test methodology.

Chapter 6 presents the architectures designed and simulated during the thesis. A short description of the idea leading up to each architecture is also given along with some brief details of the implementation.

Chapter 7 summarizes the performance of the simulated architectures. The pre-sented measurements includes: phase- and IQ-transition characteristics; Eb/n0 and ramp up ratio for different data rates. Conclusions from the

results are discussed in chapter 8.

Chapter 8 presents the conclusions made from the simulations and the overview of architectures presented in chapter 3.

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Chapter 2

Related Theory

This chapter cover some of the basic theory related to this thesis. The chapter discuss: common trade-off issues in radio frequency (RF) design related to band-width, power and data rate; frequency synthesis using charge pump based phase locked loops; minimum tone spacing in FSK systems; spectral regrowth issues and constant envelope behavior. The chapter is foremost intended for the novice RF reader.

2.1 Quadrature detection

One common QPSK receiver architecture is the quadrature detector, see figure 2.1. The received signal is down converted by a mixer using a phase aligned oscillator. Using the trigonometric function

cos(a) cos(b)= cos(a − b)+ cos(a + b)

2 (2.1)

the output from the upper mixer in figure 2.1 will be

cos(ωt + ϕ) cos(ωt) = cos([ωt + ϕ] − [ωt]) + cos([ωt + ϕ] + [ωt]) 2

= cos(ϕ) + cos(2ωt + ϕ) 2

(2.2)

and in the same way the output from the lower mixer will be cos(ωt + ϕ) cos(ωt + π/2) = cos(ϕ + π/2) + cos(2ωt + ϕ + π/2)

2 = sin(ϕ) + sin(2ωt + ϕ)

2

(2.3)

The output from the mixers hence contain one high frequency component and one DC component proportional to the phase of the incoming signal. Since

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6 Related Theory

in phase-modulation, the data is carried in the phase of the transmitted signal the high frequency component is filtered out with a low pass filter. The DC component is then integrated over one symbol time to filter out noise added by the channel before it is sampled and converted into a digital value.

LO

+90°

I {0,1}

Q {0,1}

Reset Sample&hold

Figure 2.1: Quadrature detector block diagram.

IQ diagram

IQ diagrams can be useful when evaluating radio systems and will be used in this thesis to evaluate the simulated architectures. An IQ diagram consist of the I and Q values for the demodulated signal. The I value is typically represented by the x-axis and the Q value by the y-axis. If the signal is sampled at discrete symbol times, the diagram will contain the constellation diagram in figure 2.2 (a). If the signal is sampled and plotted continuously the IQ diagram will also contain the transitions between the constellation points as in figure 2.2 (b).

The diagram can show if the transmitter and receiver are not phase aligned as in figure 2.2 (c) and can help when aligning them. IQ diagrams can also be used to view other modulation schemes beside QPSK, see figure 2.2 (d) where the diagram contain the constellation of 16-QAM. It is also possible to view the effect of gain compression in the constellation, see figure 2.2 (e).

(a) (b) (c) (d) (e)

Figure 2.2: Examples of IQ-diagram. a) Constellation diagram, b) Transition diagram, c) Constellation diagram with phase shift, d) Constellation diagram of 16-QAM and e) Constellation diagram of 16-QAM with gain compression.

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2.2 Spectrum, Power and Data Rate 7

2.2 Spectrum, Power and Data Rate

The performance of radio communication systems are limited by the available frequency spectrum, power supply and the data rate. These three components are interdependent, hence improving one parameter results in the deterioration of one or both of the other parameters. Maintaining this interdependency without violating any given specification is a major design challenge in radio communi-cation systems.

Data rate Bandwidth

Power

Figure 2.3: Illustration of the interdependency between power, bandwidth and data rate.

The majority of the power consumption in radio transmitters are usually caused by the operation of the power amplifier. This is not necessarily the case in ultra-low power transmitters since the relative output power is very low1

compared to common transmitters at higher power levels. The global efficiency deteriorates at these lower signal powers due to the frequency generation and modulation overhead starts to compete with power amplifier dissipation. For this reason special care has to be taken in the design of the frequency synthe-sis and base band modulation in order to maintain an “ultra” low global power consumption. Hence common competitive transmitter architectures like e.g. the Quadrature Raised Cosine architecture depicted in figure 2.4 are likely to suffer from poor global efficiency when used in ultra-low power applications due to its reliance on linear mixers.

2.2.1 Power and Bit Error Rate

The probability of bit error is directly determined by the signal-to-noise ratio (SNR). A larger signal-to-noise ratio reduces the error probability and vice versa.

SNR= Signal Power Noise Power =

S

N (2.4)

The error probability is however usually not expressed in terms of signal-to-noise ratio but in terms of energy per bit to the single ended thermal signal-to-noise density i.e. Eb/n0. The relation between Eb/n0and SNR is given by equation (2.5) [7].

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8 Related Theory LO +90° Spectrum Analyzer I: {-1,1} Q: {-1,1} Sine Cosine delay

Figure 2.4: Quadrature PSK architecture using Raised Cosine pulse shaping.

Eb n0 = STb n0 = SB Rbn0B = S N B Rb (2.5) Where Tbis the bit time, Rb= 1/Tbis the bit rate and B denotes the bandwidth

in Hz.

The bit error probability for PSK is given by equation (2.6) [7, 6]. Most PSK modulation schemes requires coherent (phase aligned) detectors2_{. Equation (2.6)}

is true for coherent PSK detection3_.

Pe_PSKcoherent ' erfc

r 2Eb

n0

!

(2.6) Bit error probability for coherent BFSK is given by equation (2.7) [6] and equation (2.8) [6] gives the bit error probability for non-coherent BFSK.

Pe_BFSKcoherent ' 1 2erfc r Eb 2 (2.7) Pe_BPFSKnon−coherent ' 1 2exp −Es 2 (2.8) Non-coherent detection is often used in FSK applications since it require a less complex receiver. However, the required signal power for coherent detection is lower than for non-coherent detection. Coherent BFSK requires approximately 1.5 dB less signal power than non-coherent BFSK for any given bit error rate (BER). The signal power could be reduced even further with BPSK modulation. The total reduction using BPSK is approximately 4.5 dB compared to non-coherent BFSK and approximately 3 dB compared to coherent BFSK. The bit error rates are plotted as a function of Eb/n0in figure 2.5. Quadrature PSK is a commonly used

form of PSK modulation4_{. It has a slightly worse SNR but offers higher data rates.}

2_{Some differential PSK schemes are demodulated by non-coherent detectors [7].} 3_{All error probabilities in this chapter are calculated for AWGN type channels.} 4_{Bit error probability for QPSK is given in equation 2.14.}

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2.2 Spectrum, Power and Data Rate 9

QPSK needs an Eb/n0ratio of 8.46 dB when BPSK has an Eb/n0ratio of 8 dB in

order to maintain the same BER. The difference in required SNR between BPSK and QPSK decreases as the SNR increases.

2 4 6 8 10 12 14 10−8 10−6 10−4 10−2 Eb/no (dB)

Bit Error Rate

BPSK QPSK Coherent FSK Non−coherent FSK

Figure 2.5: BER for BPSK, QPSK, coherent BFSK and non-coherent BFSK as a function of Eb/n0

2.2.2 Theoretical limits on data rate

In most cases higher data rates also means a higher bit error rate and wider bandwidth. Hence, bandwidth and bit error rate limits the data rate. In the section below about the Shannon Capacity Theorem the theoretical limit on data rate is discussed in terms of bandwidth efficiency. A discussion on how data rate affects the bit error rate for FSK and PSK modulation is presented in the section below about maximum data rates.

Shannon Capacity Theorem

The bandwidth efficiency is a major criteria when selecting modulation schemes. Bandwidth efficiency is measured in bps/Hz and is defined by equation (2.9) [7]

ηBW =

RB

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, where RB is the bit rate transmitted over an AWGN channel and BW is the

channel bandwidth in Hz. The Shannon Capacity Theorem gives the theoretical maximum capacity, i.e. the maximum bandwidth efficiency, of a channel for a given bandwidth and signal-to-noise ratio. The theorem is given in equation (2.10) [7] ηBWmax= C BW = log2 1+ Signal Power Noise Power ! bps/Hz (2.10)

, where C is the channel capacity in bps. C is also often referred to as the Shannon Limit.

Maximum data rates

Determining the relationship between data rate and bandwidth could lead to a quite lengthy discussion about the definition of bandwidth and how to appropri-ately address base band filtering techniques. The following equations (2.11) and (2.12) are presented without any further discussion on these issues. The equa-tions provides a good basis for discussion about the trade-offs of multi-valued modulation schemes [21]. RB_MFSK BT = log2(M) (1+ r)M (2.11) RB_MPSK BT = log2(M) 1+ r (2.12)

Equations (2.11) and (2.12) give the bandwidth efficiency for multivalued FSK and PSK [21]. M denotes the number of symbols used in the modulation scheme and each symbol represents log2(M) number of bits. BT denotes the transmission

bandwidth in Hz, RBis the bit rate in bps and r is a constant related to the filtering

technique and is typically in the range between 0 and 1.

Describing bandwidth efficiency as a function of the number of symbols (M) used in the modulation scheme gives opposite behavior for FSK and PSK mod-ulation. The bandwidth efficiency for PSK modulation is improved when M is increased while the bandwidth efficiency is decreased for FSK modulation. De-scribing the bit error probability as a function of the number of symbols also gives opposite behavior for PSK and FSK.

Combining equations (2.7) and (2.13) with figure 2.6 (a) visualizes that the FSK bit error rate is improved when the number of symbols (M) is increased. Equations (2.6), (2.14) and (2.15) gives the behavior of the bit error rate for PSK modulation schemes illustrated in figure 2.6 (b). The bit error rate for PSK modulation is increased when the number of symbols (M) is increased.

Pe_MFSK' M − 1 2 erfc s Eblog2(M) 2n0 , M > 2 (2.13)

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2.3 Phase Locked Loop 11 2 4 6 8 10 12 14 10−7 10−6 10−5 10−4 10−3 10−2 10−1 100 Eb/no (dB)

Bit Error Rate

Bit error probability

2 FSK 4 FSK 8 FSK (a) Multilevel FSK (MFSK) 2 4 6 8 10 12 14 10−7 10−6 10−5 10−4 10−3 10−2 10−1 100 Eb/no (dB)

Bit Error Rate

Bit error probability

BPSK QPSK 8−PSK

(b) Multilevel PSK (MPSK)

Figure 2.6: BER for multi-valued FSK and PSK

Pe_QPSK' erfc r Eb n0 " 1 −1 4erfc r Eb n0 # (2.14) Pe_MPSK ' erfc s Eblog2(M) n0 , M > 4 (2.15)

To conclude this discussion, PSK architectures can improve the bandwidth efficiency by increasing the number of symbols, at the expense of a deteriorated bit error probability. In contrast, when increasing the number of symbols used by a FSK architecture, the bandwidth efficiency is deteriorated while the bit error rate is improved.

2.3 Phase Locked Loop

Phase locked loops are a common type of architecture in frequency synthesizers. Phase locked loops are able to lock the output frequency to a reference frequency by using a negative feedback loop. The ratio between the output frequency and the reference frequency is determined by placing a frequency divider in the feed-back loop, thus providing a frequency synthesizer. A phase-frequency detector compares the divided frequency with the reference frequency and regulates the voltage level at the input of the voltage controlled oscillator. A stable and accurate output frequency is often achieved by using a crystal oscillator for the reference frequency.

There are several types of frequency dividers. The simplest type is the integer-N divider that divides the output frequency by an integer N. Frequency

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modulation can be achieved by altering the division ratio of the frequency di-vider5_.

A second order PLL is characterized by its use of a first order low pass filter at the input to the VCO. Second order indicates that the closed loop transfer function is of second order.

Phase discontinuities during FSK and PSK modulation are unwanted since it causes spectral regrowth. This is however not a major concern when using a PLL for modulation since the phase of the signal at the VCO output is always continuous.

VCO

PFD

Div.

R_p C_p C2 I_p I_p f_ref

Figure 2.7: Second order PLL with charge pump.

A closed loop transfer function of a second order PLL can be expressed as in equation (2.16).

HCL(s)=

K(s+ α)

s2_{+ Ks + α} (2.16)

An advantage using second order PLLs is the similarity to other common physical systems. Basic knowledge from control theory can be used to express the transfer function in terms of natural frequency ωnand dampening factor ζ as

can be seen in equation (2.17).

HCL(s)=

2ζωn(s+ω_2ζn)

s2+ 2ζω n+ ω2n

(2.17) The following two equations hold for the second order PLL depicted in figure 2.7 [17].

5_{There are of course more ways to produce frequency modulation with an PLL not discussed in}

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2.4 Minimum Tone Spacing for Orthogonal FSK 13 ωn= s Ip 2πCP KVCO M (2.18) ζ= Rp 2 r IpCp 2π KVCO M (2.19)

The 3 dB cut-off frequency can be expressed in ζ and ωnas follows in equation

(2.20). The equation can be derived from (2.17), (2.18) and (2.19).

f3dB= ωn s 2ζ2_{+ 1 ± 2ζ} r ζ2_{+ 1 +} 1 2ζ2 , 2π (2.20)

The step response is proportional to the inverse exponential of ζωnand can be

expressed as in equation (2.21). The equation is derived by applying a unit step to the closed loop transfer function in (2.17)

fresponse(t)= ∆ω 1 − e−ζωnt " cos(ωn p 1 − ζ2_{t) −} _√ ζ 1 − ζ2sin(ωn p 1 − ζ2_t) #! (2.21) The cut-off frequency and frequency response gives the main properties of a PLL. The maximum symbol rate is limited by the cut off frequency in the case where the modulation occurs within the frequency loop. A higher cut-off frequency enables higher symbol rate but with the cost of a higher settling time. The settling time is hence dependent on the cut-off frequency (2.21) (2.20). A short settling time is generally preferred. Short settling time often results in higher power consumption due to increased charge pump currents (2.19).

2.4 Minimum Tone Spacing for Orthogonal FSK

The maximum data rate in orthogonal FSK communication systems is limited by the ”minimum tone spacing“, also known as ”minimum frequency separation“. Since the data rate is of great importance a discussion on minimum tone spacing is presented in this section. First a mathematical view of coherent and non-coherent FSK is given and then an alternative and a perhaps more intuitive way of describing minimum frequency separation for non-coherent FSK is presented.

2.4.1 Orthogonal FSK and Coherency

As indicated in section 2.2.1, detection and demodulation of M FSK signals may be accomplished phase-coherently or non phase-coherently.

Consider the sinusoids cos(2π f1t+ φ) and cos(2π f2t). The phase φ is an

arbitrary constant angle at the interval 0 to 2π. These sinusoids are orthogonal if their convolution, equation (2.22), equates to zero.

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T

Z

0

cos(2π f1t+ φ) cos(2π f2t)dt (2.22)

Integrating and applying the limits to equation (2.22) simplifies to equation (2.23) assuming f1> f2. T is the symbol duration in seconds.

cos φ" sin 2π( f_{2π( f} 1+ f2)T 1+ f2) + sin 2π( f1− f2)T 2π( f1− f2)T # + sin φ" cos 2π( f1+ f2)T − 1 2π( f1+ f2) + cos 2π( f1− f2)T − 1 2π( f1− f2) # = 0 (2.23)

The following approximation can be done assuming f1+ f2>> 1:

sin 2π( f1+ f2)T

2π( f1+ f2)

≈ cos 2π( f1+ f2)T 2π( f1+ f2)

≈ 0 (2.24)

Combining (2.23) and (2.24) gives

cos φ sin 2π( f1− f2)T+ sin φ[cos 2π( f1− f2)T − 1] ≈ 0 (2.25)

In the non-coherent case the phase φ can assume an arbitrary value from 0 to 2π. This means that in order for the sum (of equation (2.25)) to equate to zero the terms sin 2π( f1− f2)T and cos 2π( f1− f2)T − 1 also have to equate to zero. This

gives the following equivalence.

2π( f1− f2)T= 2kπ ⇔ f1− f2= k

T (2.26)

Hence, for non-coherent minimum frequency spacing k = 1 and f1− f2 = Tk.

However in the coherent case the phase φ is known which makes it possible for the receiver to phase-align itself with the incoming signal. Since the phase is known the tone spacing for orthogonality is found in equation (2.22) by setting φ= 0, which gives

sin2π( f1− f2)T= 0 ⇔ f1− f2= n

2T (2.27)

Thus the minimum frequency separation for coherent FSK signaling occurs when n= 1 as in

f1− f2=

1

2T (2.28)

Concluding the results of the above equations, the coherent detected FSK can, for a given symbol rate, occupy less bandwidth than a non-coherent detected FSK and still be orthogonal. Remember that orthogonal FSK benefits from an optimal bit error rate for any given signal to noise ratio.

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2.5 Spectral Regrowth Issues 15

2.4.2 Alternative Approach to non-Coherent Orthogonal FSK

When modulating a FSK signal by switching between the available frequencies the individual tones will assume the shape of a sinc according to Fourier transform theory. This is visualized in figure 2.8 where the frequency spectra of binary, two tone, FSK signaling is depicted.

For a detected non-coherent tone to manifest a maximum output the peak of the tones in the corresponding frequency spectra has to align with a zero crossing of the adjacent tones. The distance from the peak of the tones main lobe and its first zero crossing gives the minimum frequency separation. That is the smallest possible separation between adjacent tones for which orthogonality is fulfilled.

Maximum bandwidth efficiency for non-coherent FSK is achieved by aligning the centers of the main lobes with the first zero crossing of the neighboring tone or tones. The minimum frequency separation for non-coherent FSK is thereby 1/T Hz where T is the symbol duration. The tones in figure 2.8 are separated with the “minimum frequency separation” distance and the highest bandwidth efficiency is therefore accomplished6_{. The tones are orthogonal, which means that}

the detected signal manifests a maximum output. Orthogonality between tones gives an optimal bit error rate for any given signal to noise ratio.

1/T Hz

f₂ f₁

T sinc(f-f₂)T T sinc(f-f₁)T

f

Figure 2.8: Minimum frequency separation for non-coherent FSK signaling. The required bandwidth of binary FSK can be derived from figure 2.8 as the minimum frequency spacing distance between the two tones plus one half of the tone spacing on both sides of the spectra. Hence the required bandwidth for binary FSK is 2/T Hz. The required bandwidth for M-ary FSK can be derived analogous to M/T Hz.

2.5 Spectral Regrowth Issues

Spectral regrowth is caused by abrupt phase changes of the transmitted signal. Remembering Fourier Series Theory, a signal with abrupt phase transitions

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cludes a large number of high frequency components. These high frequency components cause a higher percentage of the transmitted power to occur outside the designated frequency band leading to poor spectral efficiency. Base band filtering is often used to mitigate this effect by smoothering the transitions in the time domain as illustrated in figure 2.9.

LPF

Modulator

PA

x_BB(t)

Figure 2.9: Low-pass time domain filtering of the baseband signal. Filtering phase transitions reduces spectral regrowth and hence improves spec-tral efficiency. However, filtering also causes greater envelope variations, e.i. amplitude variations, of the transmitted carrier (see figure 2.10). The amplitude variations increase as the filter narrows. The operation of the power amplifier becomes a key factor in order to maintain the desired spectrum to the limited band-width. The PA must be able to follow the amplitude variations without distorting the signal by adding frequency components to the spectra. This implies that the PA needs to be linear to some degree. Larger amplitude variations requires higher linearity. The effect when a non-linear component distorts the shape of a filtered signal and deteriorates the limited bandwidth is called “spectral regrowth”.

180° 180° 90° Unfiltered QPSK Filtered QPSK t

Figure 2.10: Unfiltered and filtered QPSK waveforms.

Unfortunately, linear PAs are typically less efficient than their non-linear coun-terpart. The power efficiency of efficient linear PAs ranges about 40% and about 60% for non-linear PAs [16]. These figures are usually significantly lower for ultra low power PAs.

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2.6 Constant Envelope 17

2.6 Constant Envelope

As stated in the previous section, the term envelop is referring to the amplitude of a signal. In this section, envelope refers to the amplitude of the carrier wave signal specifically. Constant envelope simply states that the amplitude of the carrier wave signal is constant.

2.6.1 Frequency Modulation

Analog FM and discrete FSK are both constant envelope. While analog FM is by nature phase continuous, this is not the case with FSK when considering a switch based architecture (e.g. the architecture in figure 2.13 b). FSK modulation causes phase discontinuities if no consideration is taken in respect to carrier wave frequency and data rate.

FM (Analog)

Figure 2.11: Analog frequency modulation waveform.

Phase discontinuities can be avoided by choosing the difference of the carrier frequencies to be multiples of 1/2T7 _{where T is the symbol duration. Hence,}

equation (2.29) [9] must hold f1− f2 = N 2T (2.29)

, where N is an integer. Choosing the carrier frequency difference for binary FSK to the “minimum frequency separation” 1/2T discussed earlier results in a modulation scheme called Minimum Shift Keying (MSK). The phase trajectory of MSK is linear and hence also continuous. In order for the phase trajectory to also be “smooth”, its derivative also needs to be continuous which is not the case for MSK. Baseband filtering is often introduced to “smooth” the MSK signal. A common way of performing base band filtering on MSK is to introduce Gaussian filtering. Gaussian MSK or GMSK are very popular and commonly used in the industry and are found in e.g. GSM, Bluetooth and IEEE 802.11 devices.

Another way of dealing with the issues with the continuous phase during FSK modulation is to use a different type of architecture than the switching type discussed so far. By using a VCO based type of architecture, see figure 2.13 (a), a continuous phase can be guaranteed independently of carrier frequency and the data rate. The carrier frequency can be modulated by for example altering the impedance of the ocillator resonant tank.

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0

1

0

Discontinuous FSK Continuous FSK Message

Figure 2.12: Discontinuous and continuous FSK waveforms

C_Mod

Data

(a) VCO based FSK

f₁ f₂

Data (b) Switching based FSK

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2.6 Constant Envelope 19

2.6.2 Phase Modulation

PSK modulation is not regarded as being constant envelope. However, since the envelope behavior is proportional to the amount of phase discontinuity, different PSK modulation schemes exhibit different envelope characteristics.

The envelope behavior of a modulation scheme is captured by its constellation diagram. The lines between symbol states represents possible symbol transition trajectories. The envelope behavior is determined by the distance from the trajec-tory to the origin. QPSK has the largest phase shifts of the constellation diagrams depicted in figure 2.14 with a maximum phase shift of 180 degrees. A 180 degree phase transition is represented by a symbol trajectory crossing the origin. There-for, QPSK also displays the largest amplitude variations and hence the poorest envelope characteristics. π/4-DQPSK has a maximum phase shift of 135 degrees and its symbol trajectories never crosses the origin which means that it has less amplitude variations than QPSK.

QPSK π/4-DQPSK OQPSK MSK

Figure 2.14: Constellation diagrams of QPSK,π/4-DQPSK, OQPSK and MSK In OQPSK in-phase and quadrature transitions are offset by half of a symbol period which reduces the maximum phase shift to 90 degrees compared to 180 degree shift for QPSK. For constant envelope behavior the distance between the origin and the trajectory must be constant, like in the case of MSK. MSK was described in section 2.6.1 as a FSK type of modulation scheme. One other way to view MSK modulation is to view it as a PSK modulation scheme with a constant phase shift.

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Chapter 3

A Brief Overview of Current

Architectures

This chapter presents an overview of ten low-power transmitters, a description of the three most power-efficient architectures and a short discussion on using 65 nm technology in MICS applications.

3.1 A Brief Overview

An overview of existing ultra low-power transmitters and their characteristic features is presented in table 3.2 where the transmitters are ordered by their global power consumption. The transmitters are in table 3.3 ordered by the energy dissipated per bit. Some of the architectures use conventional power amplifiers with specified power efficiencies. These architectures and their efficiencies are listed in table 3.4.

The four most power efficient transmitters in table 3.3 represents three different types of architectures. A short summary of these architectures is given in the following subsections.

3.1.1 Distributed Frequency Correction

For implantable devices the MICS standard, paragraph §95.6281, specifies the frequency stability to be maintained for ±100 ppm over a range of 25°C to 45°C. The ±100 ppm requirement is relatively relaxed compared to other standards since the temperature of an implanted transmitter is constantly moderated by the human body. The architectures in [8] and [10] have taken advantage of the relaxed frequency stability requirements and completely removed the on-chip frequency loop back used for frequency correction. The frequency loop back is instead distributed to the base station, now responsible for tracking the frequency errors

1_{See section 4.1.1}

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22 A Brief Overview of Current Architectures

and periodically sending frequency correction bits to the implantable device. Both architectures utilize a digital controlled oscillator (DCO) for frequency synthesis and modulation. No crystal oscillator is needed. The frequency is controlled by capacitor banks and both architectures employs FSK modulation.

C_Array C_Mod DCO PA Data Message Message + Correction bits

Figure 3.1: DCO based transmitter with distributed frequency correction. The benefits of using a distributed frequency feedback could be questioned since many applications would need to utilize a reference frequency for other parts of the application than the transmitter. In that case the frequency loop back overhead would have a less significant footprint on the global power consump-tion.

3.1.2 Frequency Multiplying Power Amplifier

In order to improve the global power consumption [18] presents an architecture that operates entirely on the crystal frequency at 45.454 MHz. Basic building blocks are the crystal oscillator circuit, digital delay loop and the power amplifier. FSK modulation is performed at the crystal frequency by pulling the frequency with a capacitor. The crystal frequency is then forwarded to a 9-state DLL con-trolled by two feedback loops for frequency and duty-cycle control, respectively. The power amplifier operates as an edge combiner, which combines the 9 edges from the DLL producing a modulated output radio frequency at 9 × fXTAL. This

topology offers crystal stability without the need of a PLL or DLL operating at radio frequencies. One major drawback is however the lack of channel selectivity.

3.1.3 Direct VCO Modulation Using Low Supply Voltage

Paper [4] presents a transceiver architecture for wireless sensor networks and not specifically for use in MICS applications. The transmitter is designed to operate in the 2.4 GHz ISM band. The transmitter is of a direct VCO modulation type and is designed for 400 mV supply voltage to enable it to be driven from a single solar cell. Frequency control is achieved by setting a 17 bit capacitor array in the VCO. Contributors to low power consumption are: the ability to operate at a low power supply (400 mV), stacked topology in order to reuse bias current and direct VCO modulation. Further contribution is made by lowering the frequency at the

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3.2 65 nm Designs 23 CP + LF1 PDF CP + LF2 PDF 9xf_XTAL f_XTAL 45.545 MHz FSK Data A₁ A₂ A₉ Delay Chain Edge Combiner

Figure 3.2: Transmitter based on a frequency multiplying edge combiner.

counter input by preceding the counter with a dynamic ring divider. Custom dynamic logic is also used in early high frequency parts of the counter.

PA /8 Counter FLL N Tx bits 32 kHz Oscillator 50 Ohm Antenna Quadrature VCO 17 bits To the receiver

Figure 3.3: Direct VCO modulated transmitter.

3.2 65 nm Designs

The papers covered in table 3.2 and 3.3 use technologies from 180 to 90 nm. No papers were found on transmitters using 65 nm or newer technologies during the thesis. However there is at least one paper [22] on a MICS receiver utilizing the 65 nm technology. The receiver uses linear mixers and a VCO operating at radio frequency which typically would lead to high power consumption (the architecture is depicted in figure 3.4). However ultra-low power consumption is achieved by a wide use of sub-threshold devices. 90% of the transistors in all analog building blocks are operating in deep week inversion region.

Sub-threshold design has historically been associated with low frequency de-vices. Sub-threshold device operation exhibit higher transconductance but with a worsened frequency ability limiting the operation to lower frequencies. Fortu-nately, the transit frequency is increased by 75 to 100% in all operation regions for each new generation of scaling [15]. The degree of sub-threshold operation or inversion can be approximated by the inversion constant in equation (3.1).

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IC= ID I0W_L

(3.1) This 65 nm device covered in [22] is driven to deep week inversion which means that the inversion constant is less than 0.1. This in turn means that the highest power efficiency (gm/ID) is achieved for this receiver during operation at

400 MHz.

An overview of the power consumption is given in table 3.1.

PLL 900 LNA IF amp IF amp Real BPF Complex BPF Channel Selection On chip MEMS resonator RF in FSK Demod

Figure 3.4: 65 nm weak inversion MICS receiver.

Building Blocks Power Consumption (µW)

LNA 370 Quadrature Mixers 240 Complex IF BPF 500 Real IF BPF 95 IF Gain Stage 1.6 BFSK Demodulator 8 VCO 210 PLL 160

Total (Quadrature channel+ Complex IF BPF) 1490 Total (Inphase channel only+ Real IF BPF) 925

Table 3.1: Summary of power consumption.

3.3 Concluding Summary

All transmitters covered in section 3.1 are utilizing FSK modulation and are using capacitor banks for modulation and frequency selection. It is hard, if at all possi-ble2_{, to perform PSK modulation by pulling capacitors in order to alter the phase}

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3.3 Concluding Summary 25

of the signal using proposed types of architectures. Hence, modifications of the above mentioned architectures are required for use in PSK applications.

Even though smaller modifications easily could be made to generate multiple phases for PSK modulation, these architectures does not facilitate any means of filtering or smothering of the phase transitions, decremental to the overall performance.

The architectures presented in table 3.2 offer a wide range of characteristics. The data rate ranges from 50 kbit/s to 1 Mbit/s, output signal power ranges between −16 and 0 dBm while the power efficiency lies between 13 and 44%.

Categorization by Total Transmitter Power Consumption

Paper Power (Tx) Transmitter Receiver

5

0

µ

W

A 350µW CMOS MSK Transmitter and 400µW OOK Super-Regenerative Receiver for Medical Implant Communications [8]

350 µW @ -16 dBm, 120 kbit/s

DCO based with distributed feedback. No external crystal.

DCO based. Super regenerative OOK demodulation. A 490µW Fully MICS Compatible FSK

Transceiver for Implantable Devices [10]

400 µW @ -16 dBm, 250 kbit/s

DCO based with distributed feedback. No external crystal.

DCO based with relaxation mixer, Q-enhanced low-IF FSK receiver. A 500µW Neural Tag with 2µVrms AFE and

Frequency-Multiplying MICS/ISM FSK Transmitter [18] 500 µW @ -16 dBm, 100 kbit/s 9x frequency multiplying power ampliﬁer. (Edge combiner) - 5 0 0 µ W to 1 mW

An Ultra-Low Power 2.4GHz RF Transceiver for Wireless Sensor Networks in 0.13µm CMOS with 400mV Supply and an Integrated Passive RX Front-End [4] 700 µW @ -8.5 dBm, 300 kbit/s Direct FSK quadrature VCO modulation, PLL based. Single phase or quadrature down conversion; passive mixer; PLL based; FSK demodulation. 1 t o 5 m

W A 1V Wireless Transceiver for an Ultra-Low-_{Power SoC for Biotelemetry Applications [3]}

2.4 mW @ -10 dBm, 50 kbit/s Sliding-IF; PLL based; FSK and GFSK modulation. Sliding-IF; PLL based; FSK and GFSK demodulation. A 2mW 400MHz RF Transceiver SoC in 0.18µm CMOS Technology [13] 1.8 mW @ -12 dBm, 128 kbit/s

Zero-IF; PLL and mixer based; FSK modulation.

Zero-IF; PLL and mixer based; FSK modulation. 5 t o 1 0 mW

A 400-MHz CMOS Radio Front-End for Ultra Low-Power Medical Implantable Applications [5] 5.4 mW @ 0 dBm, 400 kbit/s Programmable integer-N PLL with VCO, 6 IQ mixers. Super regenerative OOK, PLL based with envelope detector. 1 0 to 1 5 mW

A Low-Power Asymmetrical MICS Wireless Interface and Transceiver Design for Medical Imaging [12]

12.7 mW @ -15d Bm, 524 kbit/s

Pseudo-open-loop PLL, high speed phase selector, G/FSK modulation.

Super regenerative OOK, PLL based with envelope detector. ≥ 1 5 m W A 400-MHz/900-MHz/2.4-GHz Multi-band FSK Transmitter in 0.18µm CMOS [11] 16 mW @ -12 dBm, 1 Mbit/s PLL based; 3-5 GHz VCO; Wide band, inductorless mixer; FSK modulation.

- An Ultra-Low Power, High Performance Medical

Implant Communication System (MICS) Transceiver for Implantable Devices [14]

16 mW @ -17 to -4 dBm,

400 kbit/s

PLL and mixer based, direct conversion, 2/4FSK modulation.

Direct conversion OOK PLL and mixer based receiver.

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Power Consumption Considering Data Rate

Energy per Bit Paper

1.60 nJ/bit A 490µW Fully MICS Compatible FSK Transceiver for Implantable Devices [10] ∼2.33 - 3 nJ/bit An Ultra-Low Power 2.4GHz RF Transceiver for Wireless Sensor Networks in 0.13µm

CMOS with 400mV Supply and an Integrated Passive RX Front-End [4]

2.9 nJ/bit A 350uW CMOS MSK Transmitter and 400uW OOK Super-Regenerative Receiver

for Medical Implant Communications [8]

4.0 nJ/bit A 500µW Neural Tag with 2µVrms AFE and Frequency-Multiplying MICS/ISM FSK Transmitter [18]

13.5 nJ/bit A 400-MHz CMOS Radio Front-End for Ultra Low-Power Medical Implantable Applications [5]

14 nJ/bit A 2mW 400MHz RF Transceiver SoC in 0.18um CMOS Technology [13]

16 nJ/bit A 400-MHz/900-MHz/2.4-GHz Multi-band FSK Transmitter in 0.18-µm CMOS [11] 24 nJ/bit A Low-Power Asymmetrical MICS Wireless Interface and Transceiver Design for

Medical Imaging [12]

40 nJ/bit An Ultra-Low Power, High Performance Medical Implant Communication System (MICS) Transceiver for Implantable Devices [14]

48 nJ/bit A 1V Wireless Transceiver for an Ultra-Low-Power SoC for Biotelemetry Applica-tions [3]

Table 3.3: Ten ultra-low power radio transmitters ordered by energy per bit.

PA Power Efficiency

Efficiency Paper

13% A 490µW Fully MICS Compatible FSK Transceiver for Implantable Devices [10] 16% A 500µW Neural Tag with 2µVrms AFE and Frequency-Multiplying MICS/ISM FSK

Transmitter [18]

31% A 1V Wireless Transceiver for an Ultra-Low-Power SoC for Biotelemetry Applica-tions [3]

32% A 400-MHz CMOS Radio Front-End for Ultra Low-Power Medical Implantable

Applications [5]

44% An Ultra-Low Power 2.4GHz RF Transceiver for Wireless Sensor Networks in 0.13µm CMOS with 400mV Supply and an Integrated Passive RX Front-End [4]

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Chapter 4

The MICS Standard and PSK

Bandwidth E

fficiency Issues

This chapter presents a summary of the Medical Implantable Communications Service (MICS) radio standard introduced by Federal Communications Commis-sion (FCC). It also points out bandwidth efficiency issues for PSK modulation complying to the MICS standard, which will be a major challenge throughout the thesis. The complete MICS standard can be found in [2].

4.1 MICS Standard

In 1999 the FCC introduced a new standard devoted to medical implantable services called MICS. The specified frequency band for MICS is 402-405 MHz. There were a number of reasons for choosing this specific band. One reason is that the propagation characteristics for frequencies in this band are favorable for transmission through the human body [19].

A summary of the MICS standard is given in the following subsections. The summary is divided into the corresponding paragraphs related to the MICS stan-dard and the main features of each paragraph are listed by bullet points. The reader is referred to “FCC Rules and Regulations Part 95” [2] for more detailed information. The related FCC paragraphs is stated in the corresponding subtitles.

4.1.1 MICS Transmitter from Paragraph §95.628

• Any frequencies from 402 to 405 MHz can be used (no channeling scheme exists).

• The emission bandwidth is 300 kHz and is measured at the points on either side of the carrier frequency situated 20 dB below the top value. Measure-ment resolution is 1% of the emission bandwidth (300 kHz).

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28 The MICS Standard and PSK Bandwidth Efficiency Issues

• A communications session involving a MICS device shall not use more than 300 kHz of bandwidth.

• Each transmitter shall maintain ±100 ppm frequency stability over a range of:

1. 25°C to 45°C for implanted transmitters.

2. 0°C to 55°C for programmer/controller transmitters.

4.1.2 Emission Types from Paragraph §95.631

• A MICS transmitter may transmit any emission type appropriate for com-munication. However, voice communication is not allowed.

4.1.3 Emission bandwidth from Paragraph §95.633

• Bandwidth limitations according to §95.628.

• Maximum EIRP is 25 µW (−16 dBm). See following paragraphs for mea-suring details.

4.1.4 Unwanted Radiation from Paragraph §95.635

• Emissions 250 kHz outside the MICS band shall be attenuated according to the following table.

Frequency MHz Field Strength (µV/m) Measurement Distance (m)

33-88 100 3

88-286 150 3

216-960 200 3

960 and above 500 3

Note - At band edges, the tighter limit apply

Table 4.1: Attenuation of signal 250 kHz outside the MICS band.

• The emission should be measured to at least the tenth harmonic of the highest fundamental frequency to be transmitted.

• Emissions within the MICS band 150 kHz from the center frequency shall be attenuated 20 dB bellow the transmitted output power.

• Any other frequencies shall be attenuated 20 dB bellow the transmitted power.

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4.1 MICS Standard 29

4.1.5 Maximum Transmitted Power from Paragraph §95.639

• The maximum EIRP for a MICS transmitter is 25 µW. Compliance may be determined by measuring the EIRP at 3 m. The equivalent radiated field at 3 meters for 25µW is 18.2 µV/m at an “open test area site”, or 9.6 µV/m which is equivalent to “free space”.

• Implantable transmitters shall be tested in a body/tissue-like medium. • The power radiated in any 300 kHz bandwidth shall not exceed 25 µW EIRP. • The antenna is considered as a part of the transmitter. Antenna and

trans-mitter are tested as a unit.

4.1.6 Additional Power Constraint from Paragraph §95.649

• No CB, R/C, LPRS, FRS, MICS, MURS or WMTS unit shall incorporate provisions for increasing its transmitter power to any level in excess of the limits specified in §95.639.

4.1.7 Crystal Control Requirements from Paragraph §95.651

• No crystal control is required for MICS.

4.1.8 Resulting Spectral Mask

The following spectral channel mask can be derived from the preceding specifi-cations for the in band frequencies (see figure 4.1).

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30 The MICS Standard and PSK Bandwidth Efficiency Issues

4.2 PSK Bandwidth E

fficiency Issues

The rational for considering the use of PSK instead of FSK modulation is that PSK modulation schemes, in common theory, require lower signal power1 _{than FSK}

based modulation schemes. PSK is also generally regarded as more bandwidth efficient facilitating the possibility of higher data rates. However, the bandwidth efficiency is affected by the spectral mask and the performed filtering. A higher amount of filtering is also affecting the signal to noise ratio leading to higher required signal powers.

S

pe

ct

ru

m

(

dB

)

-50 -40 -30 -20 -10 0 f_C+f_S f_C+2f_S f_C+3f_S f_C+4f_S f_C+5f_S f_C+6f_S f_C MSK QPSK

Figure 4.2: QPSK and MSK Spectrum

As described in chapter 2, MSK can be viewed as binary FSK modulation with minimal frequency separation. Comparing the MSK and QPSK frequency spectra in figure 4.2 reveals that MSK has a wider main lobe than QPSK and that the side-lobes in the QPSK spectra have a slower attenuation than the MSK spectra. The attenuation of the side-lobes in the MSK spectra is proportional to f4 _{while the}

attenuation in QPSK spectra is proportional to f2_{[20]. The main lobe is often the}

deciding factor when determining the bandwidth efficiency. However, it is not the case when using the MICS mask due to its 20 dB cut-offs. This drastically limits the unfiltered QPSK data rate since the first and second side-lobe fall within the 300 kHz channel bandwidth of the MICS mask (see figure 4.2 where the amplitude of the first and second side-lobe are attenuated less than 20 dB). During simulation with a resolution bandwidth of 3 kHz, also the third side lobe has to be included in the 300 kHz channel width. Limiting the theoretical data rate of the unfiltered QPSK signal to approximately 80 kbit/s. Other FSK modulation schemes with larger frequency separation distances than the minimum distance used by MSK have even higher attenuation of the unwanted side-lobes. Hence under the MICS mask, the data rate for FSK modulation is limited by the frequency separation

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4.2 PSK Bandwidth Efficiency Issues 31

while the data rate of QPSK signaling is limited by the influence of its side-lobes. The data rate of unfiltered MSK modulation is limited to about 230 kbit/s. Almost three times higher than the data rate using QPSK modulation.

Figure 4.3: Unfiltered QPSK spectra using RBW of 3 kHz.

To improve the PSK bandwidth efficiency and increase the data rate one could apply different filtering or shaping techniques in order to suppress the side-lobes. The competitiveness of using PSK modulation in MICS applications is determined by the ability to shape the frequency spectrum to better fit the MICS mask while minimizing the deterioration of the signal to noise ratio. Since additional hard-ware is most likely needed2it is also crucial that the added power consumption due to the additional hardware does not exceed the power saved by reducing the signal power.

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Chapter 5

Testbenches

This chapter describes the two testbenches used during simulations, the available measurements and the test methodology.

5.1 Detectors

What differs the two testbenches used during this thesis are the detectors. One testbench uses a PSK detector and is used for simulating the PSK architectures. The other testbench uses a FSK detector. The FSK testbench is used to validate the relative behavior of the PSK architectures by simulating FSK architectures.

5.1.1 PSK Detector

The detector that has been used is a coherent phase detector. The incoming radio frequency is down-converted with a phase aligned local oscillator. The product from this multiplication consists of one high frequency component and one DC component proportional to the phase of the input signal. The high frequency component is discarded by low pass filtering and only the DC component is used. The detection and demodulation are analogous for the in-phase and quadrature paths. They differ however in respect to the local oscillator which is shifted 90° between the paths.

After the multiplication and filtering, the DC component is integrated over at most a symbol period and then sampled. Different modulation techniques requires different timing. The timing for the two paths are controlled individually by an integrator reset and a sample-and-hold signal for each path respectively. The PSK demodulator is depicted in figure 5.1.

5.1.2 FSK Detector

The FSK detector used is a binary non-coherent quadrature detector. Hence, it does not track the phase of the transmitted carrier frequency. A non-coherent

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34 Testbenches LO +90° I {0,1} Q {0,1} Reset Sample&hold

Figure 5.1: PSK demodulator block diagram.

detector requires an Ebto n0ratio which is 1.5 dB higher than its coherent

equiv-alent1_.

The binary modulated FSK signal is composed of two alternating carrier fre-quencies with a frequency spacing of∆ f . The received signal is down-converted in the mixer stages which in turn are followed by integrator stages acting as low-pass filters by removing higher frequency components. The resulting DC components are then sampled and squared. If the received carrier frequency is equal to fcthe decision branch will assume a positive value while it will assume

a negative value in the case where the carrier frequency is equal to fc+ ∆ f . The

values in the decision branch are converted to a binary sequence by a comparator. The FSK detector is depicted in figure 5.2.

5.2 Measurements

To determine the performance of the architecture under test, the testbench mea-sures the spectrum of the modulated signal. Additive white Gaussian noise is added before demodulation to simulate a noisy channel. The demodulated signal is then compared with the input data to test for bit errors under a predetermined SNR. The following measurements are available in the testbench.

• MICS mask compliance • BER measurement

• Eye diagram of the integrated signal to tweak the timing • SNR measurement

• Eb/n0measurement

• Channel power and Adjacent channel power 1_{See section 2.2.1.}

High Level Ultra Low Power Transmitters for the MICS Standard

Institutionen för systemteknik

Department of Electrical Engineering

Examensarbete

A Study on QPSK Modulator Architectures for Ultra

Low Power Transmitters

A Study on QPSK Modulator Architectures for Ultra

Low Power Transmitters

Examensarbete utfört i Elektroniska komponenter

vid Tekniska högskolan i Linköping

av

Abstract

Acknowledgments

Contents

List of Tables

List of Figures

Abbreviations and Acronyms

Chapter 1

Introduction

1.1

Purpose

1.2

Background

1.3

Goal

1.4

Method

1.5

Delimitations

1.5.1

High Level Architectures

1.5.2

Mixer-Less Architectures

1.5.3

MICS Standard

1.6

Tools

1.7

Report Outline

Chapter 2

Related Theory

2.1

Quadrature detection

2.2

Spectrum, Power and Data Rate

2.2.1

Power and Bit Error Rate

2.2.2

Theoretical limits on data rate

2.3

Phase Locked Loop

VCO

PFD

Div.

2.4

Minimum Tone Spacing for Orthogonal FSK

2.4.1

Orthogonal FSK and Coherency

2.4.2

Alternative Approach to non-Coherent Orthogonal FSK

2.5

Spectral Regrowth Issues

LPF

Modulator

PA

2.6

Constant Envelope

2.6.1

Frequency Modulation

0

1

0

2.6.2

Phase Modulation

Chapter 3

A Brief Overview of Current

Architectures

3.1

A Brief Overview

3.1.1