Experimental Transient Behaviour Characterisation of Induction Motor fed by Variable Frequency Drives for Pump Applications
AMER HALILOVIC
KTH ROYAL INSTITUTE OF TECHNOLOGY
Experimental Transient Behaviour Characterisation of Induction Motor fed by Variable Frequency Drives
for Pump Applications by
Amer Halilović
Master Thesis
in Electrical Machines and Drives
Royal Institute of Technology School of Electrical Engineering
Department of Electrical Energy Conversion Supervisor: Martin Zetterquist, Xylem
Examiner: Juliette Soulard, KTH
Stockholm, Sweden, 2014
Abstract
The use of variable frequency drives in centrifugal pump applications has raised the question of how to select a drive. Clogging obstacles in waste water applications create unknown transient loads for the pump system. A sudden load increase occurrence can clog the pump if the drive cannot supply enough current to reach the motor’s torque demand.
In order to select a suitable drive, an empirical approach has been im- plemented, investigating three different drives. Results have shown that selecting a drive with the highest possible overload capabilities, even if for a short time is most suitable. Operation in vector speed control gives the most reliable operation if an automatic parameter tuning is performed by the drive.
Keywords: Variable Frequency Drive, Centrifugal Pump, Induction Motor, Volts-per-Hertz Control, Speed Control, Overdi- mensioning
Användningen av frekvensomriktare i centrifugalpumpar har väckt frågan om hur en omriktare skall väljas. Igensättande objekt i avlopps- vatten kan ge upphov till transienta laster i pumpsystemen. En oförut- sedd lastökning kan sätta igen pumpen om frekvensomriktaren inte kan förse motorn tillräckligt med ström för att möta momentbehovet. För att välja en lämplig omriktare har ett empiriskt tillvägagångssätt valts i en undersökning av tre olika omriktare. Resultat har visat att det är lämpligast att välja en omriktare med högst överbelastningskapacitet, även om under en kort tid. Vektor hastighetskontroll är metoden som ger stabil körning om omriktaren fått automatiskt ställa in motorpara- metrarna.
Nyckelord: Frekvensomriktare, centrifugalpump, asynkron- motor, skalärkontroll, hastighetskontroll, överdimensionering
Acknowledgement
I would like to thank my supervisor Martin Zetterquist, Development Engineer at Xylem Water Solutions AB along with the R&D Manager for Electrical Components and Systems Jürgen Mökander for the opportunity and the support I received. I would also like to thank my examiner at KTH, Juliette Soulard, that has been an inspirational source and guided me with feedback throughout the project. I would also like to acknowledge the participants in the reference group who contributed with valuable feedback. A special thanks goes to Per Miskas who assisted me in the lab.
List of Symbols
I Current [A]
T Torque [T]
T /I Torque-over-current-ratio [Nm/A]
f0 Frequency [Hz]
fsw Switching frequency [kHz]
Vref Reference voltage [V]
Vcarrier Carrier voltage [V]
p Pulse number [-]
M Modulation index [-]
Contents
1 Introduction 1
1.1 Background . . . 1
1.2 Aim and Scope . . . 3
1.3 Structure of the Report . . . 3
2 Theory Behind Variable Frequency Drives 5 2.1 Overview of a Variable Frequency Drive . . . 5
2.2 Power Electronics . . . 5
2.3 Output Waveforms . . . 7
2.4 Control . . . 9
2.4.1 Volts-per-Hertz Control . . . 10
2.4.2 Vector Control . . . 10
2.5 Drive Requirements . . . 11
2.6 Direct On-Line . . . 11
2.7 DC-link . . . 12
3 Tested Drives and Motor 15 3.1 Tested Drives . . . 15
3.1.1 Hardware . . . 15
3.1.2 Functionality . . . 16
3.1.3 Waveforms and Spectra - Measurements and Analysis . . . . 17
3.1.4 Waveforms and Spectra - Discussion . . . 27
3.2 Test Motor . . . 29
4 Test Bench Methodology 33 4.1 Materials: The Test Bench, Measurements and Post Analysis . . . . 33
4.2 Methodology: Test Procedures . . . 34
5 Test Bench Results 37 5.1 Drive A . . . 37
5.1.1 Locked Rotor Test . . . 37
5.1.2 Start Test . . . 40
5.1.3 Maximum Torque Test . . . 41
5.1.4 Speed Step Test . . . 42
5.1.5 Summary of Results . . . 44
5.2 Drive B . . . 45
5.2.1 Locked Rotor Test . . . 45
5.2.2 Start Test . . . 46
5.2.3 Maximum Torque Test . . . 47
5.2.4 Speed Step test . . . 48
5.2.5 Summary of Results . . . 49
5.3 Drive C . . . 50
5.3.1 Locked Rotor Test . . . 50
5.3.2 Start Test . . . 51
5.3.3 Load Test . . . 52
5.3.4 Speed Step Test . . . 53
5.3.5 Summary of Results . . . 56
5.4 Drive Comparison . . . 56
6 Analysis and Discussion 59 6.1 Test Bench Analysis and Discussion . . . 59
6.1.1 Drive Size Selection . . . 59
6.1.2 A Novel Evaluation Method . . . 59
6.1.3 The Test Results . . . 59
6.2 Discussion . . . 60
6.2.1 Drive Selection . . . 60
6.2.2 The Topic . . . 61
6.2.3 Simulations . . . 61
7 Conclusion and Future Work 63 7.1 Concluding Remarks . . . 63
7.1.1 Drive Selection Recomendation . . . 64
7.2 Future Work . . . 65
Appendices 65 A Complementary Results 67 A.1 Drive A . . . 67
A.1.1 Locked Rotor Test . . . 67
A.1.2 Start Test . . . 68
A.1.3 Maximum Torque Test . . . 69
A.1.4 Speed Step Test . . . 72
A.2 Drive B . . . 73
A.2.1 Locked Rotor Test . . . 73
A.2.2 Start Test . . . 75
A.2.3 Maximum Torque Test . . . 78
A.3.1 Locked Rotor Test . . . 83
A.3.2 Start Test . . . 85
A.3.3 Maximum Torque Test . . . 89
A.3.4 Speed Step Test . . . 91
References 93
Chapter 1
Introduction
1.1 Background
Centrifugal pumps are commonly used to transport liquid fluids with low viscosity, such as water and waste water for example. An electric motor powers a shaft- mounted impeller which allows, by the use of the liquids centrifugal force, to create a difference in pressure sucking the water through the pump, see figure 1.1. The load is typically proportional to the square of the speed in pump applications, which allows the electric motor to be connected to the three phase electricity network, also known as directly on-line, DOL. With the development of technology, Variable Fre- quency Drives, or motor drives, have opened up for the opportunity to control the speed of the motor, and hence the pump. This has led to that new desirable oper- ational demands can be met by means of controlling the motor. Since for example, the flow of the pumped liquid is linearly proportional to speed, the flow can be more efficiently regulated by controlling the speed. Previously, valves have been used to regulate the flow. By the means of speed regulation in order to alter the flow, significant percentage of energy savings can be achieved [1].
Figure 1.1. Cross-section of a centrifugal pump displaying the electric motor and the impeller [2].
Although it is beneficial to use a Variable Frequency Drive, VFD, it does not provide a perfectly engineered system for all applications. Considerable thought has to be put in the centrifugal pump selection. By having a pump design suited for a certain application, energy usage can be optimised along with preventing inter- nal damage and increasing the life span of the pump. A well defined technological process has to be outlined when selecting a pump. Parameters such as pressure, flow and temperature as well as the pumping service and the load curve should be known in order to define the working parameters of the pump, such as the head, capacity, suction and discharge pressures [3].
The most important aspect when selecting any component for a system is to be aware what it is intended to be used for. In pump applications, the power is pro- portional to the cube of the speed. Everything from impeller to electric motor can be designed to suit the demands. If the load curve is well defined, a VFD can ac- cordingly be selected to match the motor. Centrifugal pumps are usually considered to have non varying load characteristics. That is, the load at a certain operating point is usually always the same at that point [4]. However, the load is not always behaving ideally. The load can unexpectedly increase instantaneously if an object is sucked in towards the pump. When the object hits the impeller, it can either get stuck or in some cases get through, depending not only on the size of the object but also on the strength of the system. If the electric motor is strong enough it can overcome a sudden load increase as long as the required current is not too high.
An electric motor can usually be overloaded for a short period of time, perfectly to overcome such a load increase. The load will momentarily increase for the VFD as well, and a problem is encountered if the VFD cannot be overloaded as much as the motor. This raises a constraint on the VFD. In order to overcome the sudden load increase, the VFD has to be overdimensioned in regards to the connected motor in order to be able to supply enough current when the motor is running shortly in overload mode. Since price goes up with drive size, it is important to define precisely the level of overdimension for the VFD required by the application.
VFDs are still a technology under development. There are numerous hardware and software approaches to overcome encountered problems. Research papers have been addressing the issues on selecting a drive. Each manufacturer has respective guides on how to select the right drive for a certain application. Recommendations on how to take into consideration the name plate data of the motor and ambient of the VFD can be found, among others, in NEMA [5]. However, the answer to the question about how closely a VFD can be matched to a motor for pump applications with possible clogging was not found.
In this work, a method of investigating how to choose a drive for a centrifugal pump is presented. The problem is difficult to address considering all the varying aspects and system complexity. Therefore, empirical investigations of which factors that can influence the torque production at a given current in an electric motor for
1.2. AIM AND SCOPE
centrifugal pumps are carried out.
As mentioned, the load characteristics are familiar for these applications but it is a question of whether the VFD can supply enough current to overcome sudden transients. The dynamical aspect is unknown, and it is uncertain how often and to what extent transients can occur due to objects that are sucked in towards the impeller. The characteristics of how the exact load curve looks like is of crucial importance to be able to address the problem to full extent, and would be of great value if it was known. This new approach of empirically addressing the problem is expected open up for new ideas and narrow down the directions to take in order to gain a complete answer to the question.
1.2 Aim and Scope
The aim of this report is to present the investigations leading to the dimensioning of a variable frequency drive to an induction motor. Three drives were investigated, two from the same manufacturer, one with electrolytic capacitor bank and the other with a film capacitor bank. The third drive is from another manufacturer with an electrolytic capacitor bank. All drives were of same size with similar functionalities.
The functionalities such as energy optimising functions, acceleration ramp up times, voltage boosts, switching frequencies and scalar and vector speed control modulation methods were varied throughout the investigation. The test motor was a squirrel cage induction motor close to the power rating of the chosen drives.
1.3 Structure of the Report
The plan of this report is as follows. Chapter 2 describes the theory behind electric drives, from the power converters to control methods. Pump theory and theory behind induction motors and drive operated motors are as well included in chapter 2. In chapter 3, the tested drives are introduced, their properties and functionalities, along with the tested induction motor. At the end of the chapter, an analysis of the input and output waveforms of the drive is performed. The methodology of the performed tests is found in chapter 4 and the results in the following chapter, chapter 5. Finally, chapter 6 has further analysis following with discussion and concluding remarks in chapter 7.
Chapter 2
Theory Behind Variable Frequency Drives
2.1 Overview of a Variable Frequency Drive
A variable frequency drive for AC motors comprises different power electronic con- verters connected between the grid and the electrical machine. The basic principle is to convert the ac network voltage to dc in a rectifier in order to change it back to three-phase ac voltage with a variable frequency in an inverter [6], as illustrated in figure 2.1.
Figure 2.1. Simple block diagram of a variable frequency drive.
The dc-link block in figure 2.1 provides a stiff input to the inverter.
2.2 Power Electronics
Rectifiers, or ac-dc converters, typically convert a single-phase or a three-phase ac voltage to a dc voltage.
The input to an induction motor in centrifugal pumps is usually a three-phase supply since the motor commonly has three phases, and therefore the inputs and outputs of the VFD have as well three phases. Furthermore, there are a numerous advantages of using a three-phase rectifier over a single phase. A single-phase rec-
tifier has a much poorer power factor due to the high distortion in the line current.
The dc-current will as well contain more ripple in a single-phase opening for the opportunity for the three-phase to have smaller capacitors at the output in certain cases.
The investigated drives have three phases and diode rectifiers. The main differ- ence between a diode and a thyristor rectifier is that diode rectifiers do not have the possibility to change the dc output level and additionally, a thyristor rectifier can work as a line-commutated inverter [6, 7].
Inverters, or dc-ac converters, convert dc voltage into ac voltage and are com- mon for motor drives where the desire is to be able to control both frequency and magnitude of a sinusoidal output. The input of the converter is connected to dc- link and the output to the motor. Since the capacitors in the dc-link stabilises the voltage, the inverter is voltage stiff, hence said to be a Voltage Stiff Inverter, VSI. The drives tested in this report have all voltage stiff inverters. Current Stiff Inverters, CSI, have on the other hand an inductive dc link making it current stiff.
A drawback with current stiff converters is that the current cannot instantaneously change making it undesirable in certain pump applications [6, 7, 8].
A simple circuit diagram of a three-phase inverter is shown in figure 2.2.
Figure 2.2. Three-phase inverter [8].
The inverter in figure 2.2 is a 6-pulse inverter which the converters in the chosen drives are as well. A three-phase full-bridge converter will produce six pulses. If two six-pulse converters would be connected in series or parallel, a 12-pulse con- verter would be formed which can achieve higher voltage or current ratings. The 12-pulse converter would have a better harmonic performance but could be alike to having a six-pulse converter with an LC-filter connected to the output reducing the ripple in the output current. The cost of the above mentioned configurations are approximately the same, in no favour to any of the converters. 12-pulse converters
2.3. OUTPUT WAVEFORMS
are more common for power ranges above 1000 kW, such as in HVDC applications and traction drives [9].
2.3 Output Waveforms
The output level of the inverter in figure 2.2 varies as each switch pair is closed in sequence at a time. The pulses can be allowed to vary in length yielding to the techniques of Pulse-Width Modulation, PWM. One way is to compare a sinusoidal reference wave to a sawtooth, or triangular, carrier wave. The generated wave, see figure 2.3, will have a sinusoidal fundamental that can vary in amplitude and frequency.
Figure 2.3. PWM switching technique [6].
Each generated wave, using specific techniques, will have their corresponding harmonic spectra. Depending on the application, the unwanted harmonics can be eliminated or significantly reduced using different techniques [8].
Figure 2.4 shows an example of a harmonic spectra for a carrier based modula- tion method.
Figure 2.4. An example of a harmonic spectra for a carrier based modulation method where m and n are orders of carrier and sideband harmonics, respectively [10].
The pulse number, p, and the modulation index, M, are defined as follows, p= fsw
f0 (2.1)
where fsw is the switching frequency and f0 is the frequency of the carrier waveform, and
M = Vref
Vcarrier (2.2)
where Vref is the amplitude of the reference waveform while Vcarrier refers to the amplitude of the carrier waveform.
The first bar in figure 2.4 shows the fundamental at an order of 1. The peak value is in this case 0.8 p.u. and corresponds to the modulation index. The second harmonic present in the figure is the base band harmonic of the fundamental at order 3. Depending on the modulation technique, this harmonic can either be absent or even injected. The figure illustrates the different orders of harmonics as carrier and base band harmonics with indexes m and n, respectively. The first carrier harmonic, i.e. m = 1, is in this example at order 21. The first carrier corresponds to the pulse number p, referring to equation 2.1, where the order of the harmonic of the carrier harmonic is the pulse number. In this case, the switching frequency would then be:
fsw = p · f0 = 21 · 50 = 1050 Hz. (2.3) assuming that the reference frequency is 50 Hz, which implies that the funda- mental adapts this frequency. The second order carrier, which is situated at twice the order of the first, i.e. at harmonic order 42, is not present as seen in this ex- ample since only odd carrier multiples are present. Furthermore, each carrier is surrounded by side bands, where only even side bands are present around odd car- riers while odd side bands are only present around even carriers. Figure 2.4 is a
2.4. CONTROL
typical example of a sinusoidal reference waveform with a triangular carrier and a third-harmonic injection in the phase voltage.
By injecting a third harmonic in the phase voltage operation in the overmodu- lation region is allowed. Operating in the overmodulation region can give a higher fundamental output waveform. The injected harmonic will be eliminated in the line-to-line voltage, but it will let M increase up to 1.15. Other methods such as space vector modulation rather than the carrier PWM can be used to choose the zero vector placement in order to enter the overmodulation region safely in same manners. A quasi-square wave can give the highest value of M in the overmodula- tion region reaching M = 1.273 [8, 10].
Space vector modulation, SVM, is based on that the possible switch combina- tions for an inverter leads to discrete output voltages that can be represented as vectors. The reference vector is represented by the reference voltage generating through this technique a combination of state vectors in order to obtain an average output voltage [11].
The difference between PWM and SVM is that the latter uses a vector presen- tation of the voltages while the PWM technique uses carrier and reference waves in order to generate switching of the inverter. The switching sequence in SVM can be controlled manually in order to improve current ripple, minimise switching losses and eliminate common-mode voltages, making it more suitable to use over PWM in certain cases. Although, SVM is merely a vector presentation of PWM with a third-harmonic injection [12]. In PWM, the modulation strategy defines when the switching will occur. Despite that it is shown in [8] that switching in between half carrier periods improves the VSI maximum output voltage and the harmonic performance, it cannot be achieved by this technique. Therefore, SVM opens up to the opportunity of better harmonic performance and higher output voltages. SVM performs better than conventional PWM in the higher modulation index region [13].
The main benefit of SVM is that it has the additional degree of freedom of choosing the switches which can be used to improve the harmonic performance [8].
2.4 Control
There are several control strategies in order to operate a motor with a variable frequency drive. Basically, the strategies can be divided into two groups, scalar based controllers and vector based controllers. The drives of choice have a scalar Volts-per-Hertz, V/f -control, a Direct Torque Control, DTC, and a Speed Control, all sensorless. The two investigated strategies will be the V/f -control and the speed control since they are commonly used in pump applications and there is no clear benefit in controlling the torque over the speed.
2.4.1 Volts-per-Hertz Control
V/f-control is a traditional scalar method of controlling inverter-fed induction mo- tors. The magnitude of the stator voltage and the excitation frequency varies lin- early with each other. How a V/f -control works is illustrated with a block diagram in figure 2.5.
Figure 2.5. V/f -control block diagram.
The motor current is monitored in the current limit block. The value of the frequency is changed depending on the value of the current. The output goes to a block that adjusts the volts-per-hertz ratio defining the voltage magnitude sent to the voltage control block. Successively, the current to the motor is determined by the voltage control block that indicates the position of the flux with regards to the current. It is crucial that this angle is correct, otherwise the motor might not be in stable operation, as which can happen at low speeds or sporadic operations.
In order to keep the speed close to the desired speed, a block that alters the fre- quency reference when the load changes can be introduced, often referred as slip compensation [14].
2.4.2 Vector Control
Vector control is based on relations valid for dynamic states on the contrary to the steady state based relationships in scalar control. Instantaneous positions of voltage, current and flux space vectors can be controlled in addition to magnitude and frequency which was the only option in scalar control. In vector control, the angle between the voltage and current is known. The motor current is controlled by this angle that now opens up for improved torque control and low speed operation.
Torque current estimations can give a better slip approximation which gives a better control of the speed [14].
2.5. DRIVE REQUIREMENTS
In Field-Orientated Control, FOC, the motor flux and torque can be controlled independently. The knowledge of the exact orientation of the space vectors can help in overcoming control problems [14, 15]. With field-orientated control of induction machines, the torque can almost right away reach the torque demand [16, 17].
FOC can be rotor flux orientated or stator flux orientated where the main idea is to have a linear relationship between the control variables and torque. The relationship is achieved by transforming the motor equations in a coordinate system that rotates with the stator or rotor flux vectors, respectively [15]. The direct and indirect rotor flux orientated control were first introduced by Blaschke [18] and Hasse [19].
2.5 Drive Requirements
Choosing the right power electronics and modulation methods depends on the ap- plication and the choice of the motor drive. If the current rises above its ratings in the machine, the machine will heat up. An electric motor can be overloaded for short periods of time without damaging the motor. But on the other hand, the drive cannot withstand an overload in percentage as large as the motor. Therefore, the drive is more prone to overheat. Depending on the application, it can be de- sirable to overdimension the electronics in order to be able to overload the motor [6].
Furthermore, the switching frequency of the power converter can reduce the current ripple in the motor without having a large inductance in the motor. By increasing the switching frequency the losses in the converter increase linearly [6].
Considerations needs to be payed when choosing converters and modulation methods to that the machine is probably designed for a certain frequency from the supply. The stray and eddy current losses will increase with the use of converters because of the introduction of harmonics of different orders in the system. The introduced harmonics can lead to torque pulsations and can have an effect for lower frequencies. Depending on the motor design, the harmonic losses can be higher with a PWM inverter compared to a square-wave inverter [6] for example.
A VFD ramps up the frequency from 0 to the designated value. The current demand is relatively low if the slip can be held small. If the designated value is 50 Hz, as when connected Direct On-Line, the start current would be high.
2.6 Direct On-Line
Direct On-Line Start (DOL) is a starting method where the motor is directly con- nected to the three-phase electric power supply. The motor will draw a high current and operate at locked-rotor torque during the acceleration only to decrease the cur- rent drawn at reached speed. It is a much simpler method of starting a machine
but will have to have a stiff power supply as it can have an impact on the whole system. During the start up, the motor will build up heat which can lead to a break down if the motor is shut down and started repeatedly in an insufficient period of time [20].
In pump applications, where the load has quadratic characteristics, the acceler- ation time will be longer exposing the machine to a longer overheating time [21].
Another disadvantage of direct-on-line starting is the high torque oscillations in the initial stage of the starting process which reduces the life time of mechanical couplings and hence the pump [22].
2.7 DC-link
In order to provide a stiff input to the inverter in the drive system, a dc-link com- prised of capacitors is used that smooths the ripple in the dc output voltage and balances the instantaneous power difference between the input and the output [23].
The dc link is highlighted in figure 2.6.
Figure 2.6. The DC link in a variable frequency drive.
The operating conditions vary with capacitor type. Electrolytic capacitor tech- nology has been used for years and is now gradually being replaced by film capacitors that is now able to match the low cost. The foremost important parameters to con- sider in power electronic applications is the current and voltage ripple. A good output profile is of outmost importance in order to get a well performing inverter providing the desired output ac wave.
Electrolytic capacitors have more capacitance per volume compared to film ca- pacitors. The fact yields to that more film capacitors, in number, have to be used in order to match the size of the capacitance of electrolytic type. Film capacitors are therefore charged and discharged more frequently per a 20 ms period. This can create more ripple in the dc voltage. A high ripple can create difficulties in modulating the rectified voltage. One of the reason why film capacitors are not as common as electrolytic capacitors on the market.
There is a trade-off when replacing electrolytic capacitors with film. The number of film capacitors needs to increase in order to keep the ripple down to the same level and be able to remain at the same switching frequency.
2.7. DC-LINK
The benefit of using film capacitors is that they can achieve a higher power factor and a higher cosφ compared to electrolytic capacitors.
The power factor is defined as the ratio of active power over apparent power.
With the introduction of harmonics, the power factor will decrease. More current will be needed to deliver the same amount of active power in that case. Although, cosφ can still remain the same. cosφ is the angle between the voltage and the current. The difficulty is to maintain a power factor as close to 1 as possible.
Chapter 3
Tested Drives and Motor
3.1 Tested Drives
Three drives were investigated. The selection of the drives was mainly based on what was already in use with induction motors in pump applications. All three drives were of same nominal ratings. Two were from the same manufacturer with same functionalities but with different hardware, named here Drive A and Drive B, respectively. The third drive was from a different manufacturer, named Drive C in this report.
3.1.1 Hardware Drive A
Output Power: 15 kW Rated Voltage: 400 V Rated Current: 30 A
DC-link Capacitors: Electrolytic
Overload Capability: 200% for 4 seconds Drive B
Output Power: 15 kW Rated Voltage: 400 V Rated Current: 30 A DC-link Capacitors: Film
Overload Capability: 200% for 4 seconds Drive C
Output Power: 15 kW Rated Voltage: 400 V Rated Current: 30 A
DC-link Capacitors: Electrolytic
Overload Capability: 110% for 60 seconds
3.1.2 Functionality
Drive A and B, from the same manufacturer, have the same functions. Everything from available switching frequency to control methods is similar. The third drive shares some of the functions found in drive A and B. They are though designed in other manners since the drive is from another manufacturer.
Following is a list of drive parameter settings for each drive and description of their functionalities.
Automatic Parameter Tune- Drive A, B and C
The electric parameters of the connected motor are determined by initiating this function.
Energy Optimising Function- Drive A, B and C
Drive A and B’s energy optimising function works in the way that the out- put voltage applied to the motor is reduced in order to reduce the overall energy consumption. The function is intended to be used when the motor is running at constant speed and when it is not heavily loaded.
For Drive C, the function is described as to make the output voltage suit the current load situation. The nominal value of cosφ has to be set correctly in order for the function to work optimally. It does as well reduce the applied voltage at a given low load.
Acceleration Ramp Up Time - Drive A, B and C
The parameter determines how long time it will take for the frequency to accel- erate from 0 to its designated value in Hertz.
The lowest value that can be set for drive A and B is 0.0 seconds. 1 second is the lowest value for drive C.
Switching Frequency- Drive A, B and C
The switching frequency for Drive A and B can be set from 4 to 24 kHz while in drive C, from 2 to 16 kHz.
Control- Drive A and B
The control methods for Drive A and B comprises of a scalar Voltz-per-Hertz,
3.1. TESTED DRIVES
V/f-control, and a vector speed control.
V/f Voltage Boost - Drive A and B
The function increases the applied motor voltage at low frequencies. The func- tion only works in V/f -control. The purpose is to make motor starts easier.
Switching Pattern - Drive C
Two different switching patterns are available within drive C: SFAVM, Stator Flux Asynchronous Vector Modulation, and 60◦AVM, 60 Degree Asynchronous Vec- tor Modulation. 60◦ AVM is suitable for low speed performance and SFAVM suits high speed efficiency and at full motor output.
Overmodulation - Drive C
Drive C offers the possibility whether to allow the VFD to enter the overmodu- lation region or not.
Modulation Random - Drive C
Drive C has a function that reduces acoustic motor switching noise by changing the synchronism of the PWM.
3.1.3 Waveforms and Spectra - Measurements and Analysis
The following section analyses waveforms and harmonic spectra measured on the input and outputs of the three drives.
The drives were connected to induction motors with name plate data:
Motor 1:
3-phase, 50 Hz, 3.1 kW, 1445 rpm, 400 V, 6.7 A, cosφ=0.80 Motor 2:
3-phase, 50 Hz, 15 kW, 2915 rpm, 400 V, 27 A, cosφ=0.88
Both motors were run at no load, as the objective was to study the shape of the waveforms and the harmonic spectra.
The input current waveform of Drive A connected to Motor 1 is shown in figure 3.1. The drive was set to operate in V/f -control and the switching frequency was set to 4 kHz.
Figure 3.1. Drive A connected to motor 1, input phase current waveform, operated in V/f -control with fsw = 4 kHz.
The shape of the current in figure 3.1 indicates that Drive A has electrolytic capacitors in the dc-bank. The switching frequency or the modulation method did not have an impact on the waveform nor the corresponding harmonic spectrum in the input phase current.
The output phase voltage and the corresponding harmonic spectrum of drive A connected to motor 1 is shown in figure 3.2 when the drive was running in V/f - control. The switching frequency was set to 4 kHz.
3.1. TESTED DRIVES
Figure 3.2. Drive A connected to motor 1, output phase voltage waveform and corresponding harmonic spectrum, V/f -control, fsw = 4 kHz.
Figure 3.2 shows the phase voltage waveform and the corresponding spectrum.
It is visible that only odd carrier harmonics are present and only even side bands around odd carriers and odd side bands around even carriers. There is also a third harmonic present with a 1/6th amplitude of the fundamental. The switching fre- quency can be calculated by the use of equation 2.1 giving:
Drive settings: fsw = 4 kHz, calculated: fsw= p · f0= 40 · 50 = 2 kHz
When the switching frequency was later set to 16 and 24 kHz, the pulse number was at an order of 160 and 240, respectively, giving:
Drive settings: fsw = 16 kHz, calculated: fsw= p · f0= 160 · 50 = 8 kHz Drive settings: fsw = 24 kHz, calculated: fsw= p · f0= 240 · 50 = 12 kHz The above results show that the actual switching frequency is only half of the switching frequency set in the drive parameter settings.
Figure 3.3 shows the line-to-line output voltage waveform and corresponding harmonic spectrum of drive A connected to motor 1.
Figure 3.3. Drive A connected to motor 1, output line-to-line voltage waveform and corresponding harmonic spectrum, V/f -control, fsw = 4 kHz.
The third harmonic is cancelled in the line-to-line voltage shown in figure 3.3 yielding to a smaller harmonic distortion, in accordance to theory. As before, even side bands are present around the odd carrier, while odd side bands are present around even carriers. Both the first odd and the first even carrier are suppressed.
Similar to before, the pulse number in figure 3.3 is 40 giving half the switching frequency of the stated by the drive manufacturer. As the switching frequency was increased to other values, half of what was set was obtained as for previous cases.
Figure 3.4 shows the input current waveform of the film capacitor drive, Drive B, connected to motor 2.
3.1. TESTED DRIVES
Figure 3.4. Drive B connected to motor 2, input phase current waveform and corresponding harmonic spectrum, V/f -control, fsw = 4 kHz.
The waveform of the output current shown in figure 3.4 suggests that the dc- bank is of film capacitor type. The red line shows the level of the RMS value.
Drive B connected to motor 2 was running in V/f -control. The switching fre- quency was set to 4 kHz. The output waveform of voltage and corresponding har- monic spectrum is shown in figure 3.5.
Figure 3.5. Drive B connected to motor 2, output phase voltage waveform and corresponding harmonic spectrum, V/f -control, fsw = 4 kHz.
The figure, 3.5, showing the phase voltage shows similar results to as before with around odd carriers odd side bands are suppressed and around even carriers even side bands are suppressed. In this case, there was no third harmonic injection.
What is interesting to note is that the indicated pulse number, here p = 100, differs from before:
Drive was set to fsw= 4 kHz, calculated: fsw = p · f0 = 100 · 50 = 5 kHz Figure 3.6 shows the line-to-line voltage of drive B connected to motor 2.
3.1. TESTED DRIVES
Figure 3.6. Drive B connected to motor 2, output line-to-line voltage waveform and corresponding harmonic spectrum, V/f -control, fsw = 4 kHz.
The result of the switching frequency mentioned above was confirmed as well in the line-to-line output, figure 3.6.
Figure 3.7 shows the phase voltage and corresponding spectrum when the switch- ing frequency was set to 16 kHz in drive B, connected to motor 2.
Figure 3.7. Drive B connected to motor 2, output phase voltage waveform and corresponding harmonic spectrum, V/f -control, fsw = 16 kHz.
Figure 3.7 shows that now there was a third harmonic injected in this case. The calculated switching frequency from the harmonic spectrum tells:
Drive was set to fsw= 16 kHz, calculated: fsw = p · f0 = 160 · 50 = 8 kHz The actual switching frequency was only the half of what was set in the drive, at 16 kHz.
The input phase current waveform of Drive C connected to motor 2 is shown in figure 3.8. The switching frequency was set to 4 kHz and the switching pattern was 60◦ AVM.
3.1. TESTED DRIVES
Figure 3.8. Drive C connected to motor2, input phase current waveform, fsw = 4 kHz.
The current shown in figure 3.8 shows typical characteristics of electrolytic ca- pacitors in the dc-bank.
The output waveform and corresponding spectra of the phase and line-to-line voltage waveform is displayed in figure 3.9 and 3.10, respectively, for drive C con- nected to motor 2. The switching pattern was 60◦AVM and the switching frequency 2 kHz.
Figure 3.9. Drive C, output phase voltage waveform and corresponding harmonic spectrum, fsw = 2 kHz.
3.1. TESTED DRIVES
Figure 3.10. Drive C, output line-to-line voltage waveform and corresponding har- monic spectrum, fsw = 2 kHz.
Figure 3.9 and 3.10 show the phase and line-to-line voltage, respectively, when the switching frequency was set to 2 kHz. The harmonic spectra for both waveforms indicate a pulse number of 40 yielding to a switching frequency of 2 kHz. The third harmonic is higher in amplitude than the fundamental in the phase voltage but is cancelled out, as before, in the line-to-line voltage. More low order harmonics appear in the phase voltage compared to the other drives but are suppressed in the line-to-line voltage. As for previous drives, the phase voltage has only odd carrier harmonics. The odd side bands around odd carriers are suppressed while the even sidebands are suppressed around even carriers. Furthermore, in the line-to-line volt- age, the first carrier is suppressed.
3.1.4 Waveforms and Spectra - Discussion
A drive manufacturer keeps secret how the output waveforms are modulated. With that in mind, it is clear that it is quite difficult to figure out the modulation tech- niques. What can be done is to evaluate the harmonic spectrum, the harmonic distortion and to look at the waveforms, the frequency of the fundamental, if it is near what is promised, and the amplitudes. As a customer, what is interesting to
know is how to operate the drive, which functions will do what and what will be the performance. Another important thing for the user is to understand the impact of the output on their application.
The method chosen in this report to evaluate the drive performance was to look at the waveform, the shape and amplitude and the harmonic spectrum. The raw data of the output was sampled with an oscilloscope to a USB which was then post processed in Matlab in order to generate the harmonic spectrum. A few points might get lost in this process considering that the oscilloscope can only save 1000 points onto a USB-stick. Since the output frequency on the VFD’s was chosen to 50 Hz, the scope of one period on the oscilloscope was 20 ms, which is just below the boundary of possible aliasing effects, considering the fast switching frequency in the kHz-range.
The results from the output waveforms of Drive A were surprising. The switch- ing frequency indicated by the harmonic spectrum showed always half of what was configured in the drive. It is strange that there would be harmonic groups in orders lower than the first carrier group of harmonics.
Drive B, which is the same type and from the same manufacturer as Drive A but with film capacitors instead of electrolytic capacitors in the dc-bank, showed also a deviating switching frequency compared to the configured value in the drive.
For a low frequency of 4 kHz set in the drive the switching frequency was actually 1 kHz higher, while for a higher switching frequency of 16 kHz, the obtained switching frequency was 8 kHz. A 1 kHz higher switching frequency might not matter in some applications, but half the switching frequency of the promised can lead to undesired performances or unwanted noise, depending on the application.
Since both Drive A and B are from the same manufacturer it could be that they use the same modulation techniques for both drives, but since the dc bank is different the power factor might be better for one of the drives. Nevertheless, the pulse number indicated by the harmonic spectrum should still correspond to the switching frequency since it is defined in that manner.
The results obtained by Drive C, which indicate that the switching frequency configured in the drive corresponds to the obtained by the generated harmonic spectra in Matlab allows to conclude that the measurement method and the post processing is done accurately.
The harmonic spectra indicate that the modulation techniques will for instance allow an entering in the overmodulation region, as can be seen with third-harmonic injections, yielding to a higher voltage output. With the possibility of higher output values, the motor can be allowed to produce more torque.
3.2. TEST MOTOR
3.2 Test Motor
The selected motor was an induction motor with a squirrel cage rotor. The name- plate info and the motor parameters are found in table 3.1.
Table 3.1. Motor parameters and nameplate data of the induction motor.
Frequency 50 Hz Rs 0.799 Ω
Voltage 3×400 V Rr 27.3 Ω
Current 27 A Xs 1.80 Ω
No. of Phases 3 Xr 1.63 Ω
Speed 2915 rpm Xm 63.9 Ω
Power Factor 0.88
Input Power 16.6 kW Torque 49 Nm
Shaft Power 15 kW Rotor Inertia 0.019 kgm2
Theoretical calculated torque and current versus speed curves are found in fig- ures 3.11 and 3.12, respectively.
Figure 3.11. Theoretically calculated torque-speed curve according to given motor parameters in table 3.1.
Figure 3.12. Theoretically calculated current-speed curve according to given motor parameters in table 3.1.
The curves shown in figure 3.11 and 3.12 are elaborated to study a running motor. The theoretical curves are designed to replicate the behaviour from the speed at the pull-out torque, 2250 rpm, to synchronous speed, 3000 rpm. Therefore, it does not reflect the starting behaviour truly.
According to figure 3.11 and 3.12, the starting torque, at speed 0 rpm, is 105 Nm drawing 215 A. The maximum torque occurs at 2250 rpm with a current of 150 A giving 175 Nm. At a 100 % load the speed is 2915 rpm and the torque and current are 49 Nm and 27 A, respectively. At no load the current is 10 A.
A summary of the Direct On-Line, DOL, values are found in table 3.2. The values are obtained from running the motor directly connected to the grid. The shaft was connected to a test bench with a relatively high inertia compared to a centrifugal pump when started at standstill in water. When the motor is running with minimum load it has the additional inertia from the test bench. It is therefore considered to be loaded with the least possible load.
3.2. TEST MOTOR
Table 3.2. DOL performance values of the tested induction motor.
Run Voltage [V] cosφ [-] Speed [rpm] Torque [Nm] Current [A]
Locked Rotor - - 0 137.5 229.7
125% Load 400.7 0.9214 2864.6 75.01 40.24
100% Load 400.0 0.9131 2899.2 59.29 31.89
75% Load 400.9 0.8827 2928.7 44.03 24.26
50% Load 399.8 0.8114 2951.7 29.12 17.60
25% Load 399.5 0.5991 2976.4 14.45 12.24
Min. Load 398.9 0.3213 2993.5 5.36 9.96
The locked rotor test values in table 3.2 are extrapolated at rated voltage, 400 V, from tests performed at lower ratings. The tests are carried out in that way in order to prevent too high currents forces in the test bench. Values in table 3.2 is in accordance with figures 3.11 3.12 between the two points, 2250 and 3000 rpm.
Chapter 4
Test Bench Methodology
4.1 Materials: The Test Bench, Measurements and Post Analysis
In the following section the equipment used to perform the tests are listed and briefly described.
Drives and Motor The drives described in section 3.1 are the test objects and the motor described in section 3.2 acts as the load of the test objects.
Test Bench The motor, connected to the VFD, was mounted in a test bench.
The shaft was connected to a braking motor which applied a load on the motor shaft. The test bench had therefore its own inertia which the induction motor had to overcome.
Braking Motor The braking motor was a HDP Servomotor from ABB Sace - Italy. The motor had a maximum speed of 3600 rpm and the rated torque 657.6 Nm.
Oscilloscope Agilent Technologies DSO7054A Oscilloscope was used to display torque and current and to save data to a USB stick which was used to transfer data to a computer for post processing purposes.
Torque Transducer HBM T10F/FS 1000 Nm, a torque transducer was used to measure the shaft torque, connected to the oscilloscope.
Ammeter Clamp Agilent 1146A, the clamp was used to measure VFD’s output phase current, connected to the oscilloscope.
Matlab® was used to post process data and generate torque and current graphs.
4.2 Methodology: Test Procedures
A way to investigate drive behaviour in waste water applications is to study the response during transient loads. By instantly increasing the load, the short term overload capability could be investigated. The amount of load and test time pe- riod could be varied in order to examine how much and for how long the drive can withstand the excessive load. Such a test would directly correspond to a clogging scenario where excessive load is applied for a short time. However, the available test bench did not have the possibility to instantly increase the load. The possible load ramp time was long. But it could still provide a feeling of how high the torque production can be and how much current can be delivered by the drive, and for how long.
The following tests were performed in the test bench.
Locked Rotor Test
The locked rotor test had the objective of studying the current and the starting torque. The test was performed by blocking the rotor. The initial current and torque was obtained at 0 rpm.
Start Test
The test was carried out by programming the drive for a ramp from stand-still to 50 Hz. The start-up capabilities of the drives were investigated by studying how low the acceleration ramp time could be programmed yet still be able to accelerate the motor and how high the torque and current were.
Maximum Torque Test
The purpose of this test was to examine how much torque the motor produces at maximum load and how much current is drawn at that point from the VFD. The test was conducted in the way that the VFD operated the motor at 50 Hz and then the test bench brake was applied and increased to full load until the drive reached its maximum current and tripped, i.e. shuts off in order to protect itself.
Speed Step Test
The output frequency of the VFD was increased from, and to, a fixed value in order to model a step change in the load. Since it is not possible to suddenly apply a short load increase by means of the test bench torque, the loading of the motor was simulated by increasing the speed. The larger the step in frequency and the shorter the acceleration time was, the heavier the load appeared to be, that is, the greater the torque increase would be. How much torque could be produced and how much current was drawn from the VFD at that point was of interest to examine. The
4.2. METHODOLOGY: TEST PROCEDURES
test bench was inactive during the tests, only contributing to the total inertia on the motor shaft, i.e. the test was run at minimum load.
VFD configurations
The VFD configurations were changed through out the tests in order to study the effects of the different possible configurations on the drive’s performance. Modu- lation techniques, switching frequency, acceleration ramp time, energy optimising functions and additional available functions were varied.
Chapter 5
Test Bench Results
The results of the test bench are divided into three sections for the three drives, respectively. Each section contains results of tests performed with altering drive settings. At the end of each drive section, a short summary is presented. The last section contains a table that summarises chosen results that demonstrated good performance. The table acts as an aid in order to make a comparison between the drives. In Appendix A, complementary results are presented in additional tables and figures.
5.1 Drive A
The results presented in this section are obtained by operating Drive A in Locked Rotor, Start, Maximum Torque and Speed Step Tests when running in V/f and speed control and altering the VFD settings: Switching Frequency, Acceleration Ramp Time and Voltage Boost, as well as running in energy optimisation.
5.1.1 Locked Rotor Test
The highest torque value in the locked rotor test was achieved in V/f -control with a voltage boost of 2.5%. The phase current and torque are shown in figure 5.1.
0.5 1 1.5 2 2.5
−80
−60
−40
−20 0 20 40 60 80
Iv
t [s]
[A]
77
0.5 1 1.5 2 2.5
0 20 40 60 80 100 120
T
t [s]
[Nm]
133
Figure 5.1. Phase current and torque, locked rotor test, V/f -control, acceleration ramp time: 1 s, Voltage boost: 2.5 %, fsw= 24 kHz.
Figure 5.1 shows that the motor gave 133 Nm which is slightly lower than the starting torque achieved with DOL, 137.5 Nm. On the other hand, the current is significantly lower 77 A, peak, compared to 229.7 A rms.
The drive was feeding the motor with a current at only 5 Hz in the test shown in figure 5.1, although the drive was set to run in 50 Hz. The torque was also pulsating at a frequency of around 5 Hz and was damped throughout the test run until the drive tripped after 3 seconds.
Figure 5.2 shows current and torque of the test that got the highest torque-over- current ratio, 1.66, in speed control.
5.1. DRIVE A
0 0.5 1 1.5 2 2.5 3
−60
−40
−20 0 20 40 60
Iv
t [s]
[A]
61
0 0.5 1 1.5 2 2.5 3
0 20 40 60 80 100
T
t [s]
[Nm]
101
Figure 5.2. Phase current and torque, locked rotor test, speed control, acceleration ramp time: 0 s, voltage boost: 0 %, fsw= 12 kHz.
The oscillation, at only 3 Hz, of the torque was lower in amplitude in speed control mode compared to V/f -control, as seen when comparing figures 5.1 and 5.2.
The drive supplied the motor with current for over 3 seconds until it tripped.
Comparing the results between V/f -control and speed control, it is noticeable that the V/f -control configuration for highest torque achieves a 30 % higher torque with a 25 % higher current demand from the VFD compared to the speed control configuration that gave the highest torque. That is, 133 Nm was the highest torque value for 77 A in V/f -control and in speed control, the highest torque was 101 Nm for 61 A. Hence, the highest torque-over-current ratio was achieved in the V/f - control run, a ratio of 1.72. Moreover, the speed control configuration gave similar results for different combinations of acceleration time, voltage boost and switching frequency while the V/f -control could not deliver any significant torque without a ramped acceleration or a voltage boost. The energy efficiency function produced more torque for less current in speed control mode compared to the V/f -control mode.
The change in switching frequency in both control modes did not indicate an impact neither on torque production nor current demand.
The results of the performed locked rotor tests can be found in the tables of section Appendix A.1.1.
5.1.2 Start Test
The start test was conducted by starting the drive from 0 to 50 Hz with different acceleration times, boost levels and in two different control modes. The switching frequency was held constant at 12 kHz for all tests.
The results from V/f -control settings indicated that with a voltage boost the acceleration time could be reduced. The minimum possible acceleration time with a 2.5 % voltage boost was 3.2 s compared to no boost at 3.3 s. If the voltage boost was chosen to 5 % the drive could not start.
The current and torque graph of a test with an acceleration ramp time set to 3.0 s and a voltage boost of 2.5 % is shown in figure 5.3.
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−100
−50 0 50 100
Iu
t [s]
[A]
78
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−20 0 20 40 60 80
T
t [s]
[Nm]
78
Figure 5.3. Phase current and torque, start test, V/f -control, acceleration ramp time: 3.0 s, voltage boost: 2.5 %, fsw = 12 kHz.
The drive tripped in the test shown in figure 5.3 but still managed to reach 50 Hz. The drive was running for just above 3 seconds. The inertia was too large to withstand the current demand at that torque production resulting in a trip. The other test runs that managed to start the motor without tripping had all around 3.5 second runs of accelerating the motor. The torque was again oscillating at around 4 Hz, only to stabilise with the increase of speed as seen in figure 5.3.
A test run was conducted when the drive was reset to factory settings. The mode was V/f -control with no voltage boost and an acceleration ramp time set to 4.0 s. The drive has no reference values to the connected motor when reset, i.e.
motor nameplate data and parameters were not set in the drive. It still managed to start the motor if the acceleration ramp time was 4.0 seconds or above.
The drive was running the motor with a high output current for a high torque
5.1. DRIVE A
value for over 4 seconds. The drive was feeding the motor with around 30 A in 0.5 s before the motor started spinning. The torque did not oscillate with these settings.
The torque-over-current ratio was 1.32 which was 30% higher than the highest value among all start tests conducted in V/f -control.
The shortest possible acceleration ramp time in speed control without tripping the drive was 2.9 seconds. A lower value compared to the shortest time of 3.2 seconds in V/f -control with a voltage boost of 2.5%. Figure 5.4 shows the current and torque of the best performing test.
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−80
−60
−40
−20 0 20 40 60 80
Iu
t [s]
[A]
77
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−20 0 20 40 60 80
100 T
t [s]
[Nm]
86
Figure 5.4. Phase current and torque, start test, speed control, acceleration ramp time: 2.9 s, fsw = 12 kHz, factory reset.
As seen from figure 5.4 the torque is not oscillating and the drive is supplying the motor with 77 A peak value for over 2.5 seconds.
Detailed presentation of results can be found in Appendix A.1.2.
5.1.3 Maximum Torque Test
The motor was running with minimum load at 50 Hz prior the applied load increase to the point of tripping the drive.
The test was conducted in V/f -control and speed control, respectively, and for different switching frequencies.
The amount of torque produced in both V/f -control and speed control mode was around 100 Nm for all runs. It could be observed that the current stayed in a smaller spread in speed control mode compared to the V/f -control, where the current had values from 70 to 90 A not following the increase of the switching frequency. The highest drawn current in the speed control configuration was when
the energy optimisation function was active, 88 A. Otherwise, the current stayed at 82-86 A. Why the current was higher when running in energy optimisation was because the applied motor voltage was decreased. The motor was operated in field weakening. As the load increased, the drive tried to compensate for the load by maintaining the torque. Since the voltage could not increase fast enough, more current was consumed.
Figure 5.5 shows the test performed V/f -control.
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5
−80
−60
−40
−20 0 20 40 60 80
Iu
t[s]
[A]
71
−2.50 −2 −1.5 −1 −0.5 0 0.5 1 1.5
20 40 60 80 100
T
t[s]
[Nm]
100
Figure 5.5. Phase current and torque, maximum torque test performed in V/f - control with fsw = 12 kHz.
Figure 5.5 indicates a linear increase of the torque as the motor was loaded until the drive tripped.
A summary of the results obtained can be found in tables in Appendix A.1.3.
5.1.4 Speed Step Test
The speed step was conducted by changing the frequency of the VFD from an initial to a final value.
With a higher frequency step and a shorter acceleration time, the slip was larger, leading to a higher torque production and demanding more current. The shortest acceleration ramp time for steps 45-50 Hz, 40-50 Hz, 31-50 Hz and 31-40 Hz was 1.5 seconds without having the drive to trip, while for the step 45-55 Hz, the drive could withstand 1.0 seconds without tripping. The choice of switching frequency did not show to have an impact on current demand and torque production. The drive could not run if it was set below 31 Hz. The V/f -control could not maintain operation at low frequencies.
5.1. DRIVE A
Figure 5.6 shows the speed step test performed in V/f -control from 40 to 50 Hz with the acceleration ramp time set to 1.5 s.
Figure 5.6. Phase current and torque, speed step test in V/f -control, fsw= 12 kHz, acceleration time: 1.5 s, 40-50 Hz, rise time: 0.3 s.
The rise time was 0.3 seconds in the test shown in figure 5.6 the current demand was 79 A. The torque was 99 Nm.
Figure 5.7 shows a speed step from 40 to 50 Hz with the acceleration ramp time set to 1.4 s giving the rise time from 40 to 50 Hz to be 0.28 seconds.
Figure 5.7. Phase current and torque, speed step test in V/f -control, fsw= 12 kHz, acceleration time: 1.4 s, 40-50 Hz, rise time: 0.28 s.
The test shown in figure 5.7 shows that the drive trips but only after the drive reaches 50 Hz. The rotor was still slipping too much demanding a high current for
a long period of time resulting in a trip of the drive.
In speed control, the shortest acceleration ramp time was set to 0.1 seconds for all frequency steps: 45-50, 40-50, 35-50 and 45-55 Hz without forcing the drive to a trip. The torque did not exceed 90 Nm while the current reached up to 78 A. The test run that produced the most torque, 90 Nm and had the highest torque-over- current ratio is shown in figure 5.8.
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−80
−60
−40
−20 0 20 40 60 80
Iu
t [s]
[A]
74
−2.5 −2 −1.5 −1 −0.5 0 0.5 1 1.5 2 2.5
−20 0 20 40 60 80
100 T
t [s]
[Nm]
90
Figure 5.8. Phase current and torque, speed step test in speed control, fsw = 12 kHz, acceleration time: 0.1 s, 40-50 Hz.
The performed test shown in figure 5.8 showed a rapid increase in frequency.
When the drive reached 50 Hz, the torque slopes down to a steady value just before the torque dips because of the rotor exceeding synchronous speed. From both torque and current curves, it is visible that the control algorithm looks different for speed control compared to V/f -control.
The results from both V/f -control and speed control can be found in tables in Appendix A.1.4.
5.1.5 Summary of Results
Drive A showed better performance when set in speed control. The switching fre- quency did not impact the results. The four tests indicate stable operation and a reliable performance. The maximum torque obtained was around 100 Nm which was lower than in V/f -control where it reached 133 Nm. The drive could start with a shorter acceleration ramp time in speed control compared to V/f -control. The drive showed better performance when an automatic parameter tune was performed compared to a factory reset. During the speed step test in speed control, the drive
