Integration of MIMO antenna with a fully 180o differential hybrid coupler

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(1)LiU-ITN-TEK-A--11/071--SE. Integration of MIMO antenna with a fully 180º differential hybrid coupler Rajkumar Kallem Sampath Kumar Veerabomma 2011-11-11. Department of Science and Technology Linköping University SE-601 74 Norrköping , Sw eden. Institutionen för teknik och naturvetenskap Linköpings universitet 601 74 Norrköping.

(2) LiU-ITN-TEK-A--11/071--SE. Integration of MIMO antenna with a fully 180º differential hybrid coupler Examensarbete utfört i elektroteknik vid Tekniska högskolan vid Linköpings universitet. Rajkumar Kallem Sampath Kumar Veerabomma Examinator Magnus Karlsson Norrköping 2011-11-11.

(3) Upphovsrätt Detta dokument hålls tillgängligt på Internet – eller dess framtida ersättare – under en längre tid från publiceringsdatum under förutsättning att inga extraordinära omständigheter uppstår. Tillgång till dokumentet innebär tillstånd för var och en att läsa, ladda ner, skriva ut enstaka kopior för enskilt bruk och att använda det oförändrat för ickekommersiell forskning och för undervisning. Överföring av upphovsrätten vid en senare tidpunkt kan inte upphäva detta tillstånd. All annan användning av dokumentet kräver upphovsmannens medgivande. För att garantera äktheten, säkerheten och tillgängligheten finns det lösningar av teknisk och administrativ art. Upphovsmannens ideella rätt innefattar rätt att bli nämnd som upphovsman i den omfattning som god sed kräver vid användning av dokumentet på ovan beskrivna sätt samt skydd mot att dokumentet ändras eller presenteras i sådan form eller i sådant sammanhang som är kränkande för upphovsmannens litterära eller konstnärliga anseende eller egenart. För ytterligare information om Linköping University Electronic Press se förlagets hemsida http://www.ep.liu.se/ Copyright The publishers will keep this document online on the Internet - or its possible replacement - for a considerable time from the date of publication barring exceptional circumstances. The online availability of the document implies a permanent permission for anyone to read, to download, to print out single copies for your own use and to use it unchanged for any non-commercial research and educational purpose. Subsequent transfers of copyright cannot revoke this permission. All other uses of the document are conditional on the consent of the copyright owner. The publisher has taken technical and administrative measures to assure authenticity, security and accessibility. According to intellectual property law the author has the right to be mentioned when his/her work is accessed as described above and to be protected against infringement. For additional information about the Linköping University Electronic Press and its procedures for publication and for assurance of document integrity, please refer to its WWW home page: http://www.ep.liu.se/. © Rajkumar Kallem, Sampath Kumar Veerabomma.

(4) Integration of MIMO antenna with a fully differential 180o hybrid coupler. Thesis by Rajkumar Kallem Sampath Kumar Veerabomma. In the Fulfillment of the Requirements for the Degree of Master of Science. Department of Science and Technology Linköping University, SE-601 74 Norrköping, Sweden Norrköping 2011.

(5) i. Abstract This thesis work proposes the integration of MIMO antenna (i.e., two bowtie dipole antennas) with a fully differential 180o hybrid coupler in the frequency range of 0.7-3 GHz which falls in the Ultra High Frequency (UHF) band as per the spectrum allocated by International Telecommunication Union (ITU). The applications in this band are Long Term Evolution (LTE), Global System for Mobile Communication (GSM), Global Positioning System (GPS), Digital Communication System, Personal Communication System, Universal Mobile Telecommunication System (UMTS), Wireless Local Area Network (WLAN) and Global star satellite phone downlink and uplink. This thesis introduces a new technique of integrating two bowtie dipole antennas with a fully differential 180o hybrid coupler. The first part is design of bowtie dipole antenna in the desired frequency range of 0.7-3 GHz simulated using Advanced Design System (ADS) 2009 from Agilent Technologies Inc. In the design of bowtie dipole antenna started with a stepwise design with simple dipole antenna followed by parametric study. After literature study on different articles of wideband dipole antennas finally a bowtie dipole antenna was designed which covers the required bandwidth. The second part is to design a differential 180o hybrid coupler in the frequency range of 0.7-3 GHz simulated using ADS. Before implementing the actual differential 180 o hybrid coupler started with a design of fundamental single ended 90o coupler followed by single ended 180o coupler and by analyzing the results of both finally differential 180 o hybrid coupler was designed and observed the simulated results. The last part of the thesis work is to integrate MIMO antenna with a fully differential 180o hybrid coupler and to compare the results of isolation between antennas with and without coupler. To see in which degree the isolation can be improved by adding an 180o hybrid coupler to the antennas..

(6) ii. Acknowledgement Firstly we would like to thank Dr. Magnus Karlsson for giving us this opportunity to perform our master thesis in ITN Department and also for his guidance, support, encouraging attitude and also for his valuable knowledge during the semester period, throughout the completion of thesis. We would like to thank our supervisor Owais for his guidance and full assistance helped us in achieving each and every task in the thesis. He was very kind hearted who always answered our questions even when he was busy with his work, which helped us in gaining knowledge related to thesis work. We would also like to thank our parents for their prayers, love and continuous support gave us strength to complete our thesis and also we would like to thank all our friends for their encouragement and support to make this thesis possible..

(7) iii. List of Abbreviations ADS. Advanced Design System. CAD. Computer Aided Design. EDA. Electronic Design Automation. GPR. Ground Penetrating Radar. GPS. Global Positioning System. GSM. Global System for Mobile Communication. ITU. International Telecommunication Union. LTE. Long Term Evolution. MIMO. Multiple Input Multiple Output. RF. Radio Frequency. UHF. Ultra High Frequency. UMTS. Universal Mobile Telecommunication System. VSWR. Voltage Standing Wave Ratio. WLAN. Wireless Local Area Network.

(8) iv. Table of Contents Abstract.................................................................................................................................................... i Acknowledgement ................................................................................................................................... ii List of Abbreviations .............................................................................................................................. iii Table of Contents ................................................................................................................................... iv CHAPTER 1 .......................................................................................................................................... 1 1. Introduction ......................................................................................................................................... 1 1.1 Background.................................................................................................................................... 1 1.2 Objective ....................................................................................................................................... 2 1.3 Thesis overview ............................................................................................................................. 2 CHAPTER 2 .......................................................................................................................................... 3 2. Theory................................................................................................................................................. 3 2.1 Antenna ......................................................................................................................................... 3 2.1.1 Input impedance ..................................................................................................................... 3 2.1.2 Bandwidth .............................................................................................................................. 3 2.1.3 Input reflection ....................................................................................................................... 4 2.1.4 VSWR ...................................................................................................................................... 4 2.1.5 Directivity ............................................................................................................................... 4 2.1.6 Gain ........................................................................................................................................ 4 2.1.7 Radiation intensity .................................................................................................................. 5 2.1.8 Front to back ratio .................................................................................................................. 5 2.1.9 Radiation pattern .................................................................................................................... 5 2.1.10 Side lobes .............................................................................................................................. 6 2.1.11 Polarization ........................................................................................................................... 6 2.1.12 Beamwidth............................................................................................................................ 6 2.1.13 Field regions.......................................................................................................................... 6.

(9) v 2.1.14 Antenna efficiency ................................................................................................................ 7 2.1.15 Envelope correlation ............................................................................................................. 7 2.2 Dipole antenna .............................................................................................................................. 8 2.2.1 Antenna length ....................................................................................................................... 8 2.2.2 Dipole feed impedance ........................................................................................................... 9 2.2.3 Wideband dipoles ................................................................................................................... 9 2.2.4 Parasitic element .................................................................................................................. 10 2.3 Hybrid coupler ............................................................................................................................. 10 2.3.1 90° Hybrid coupler ................................................................................................................ 10 2.3.2 180° Hybrid coupler .............................................................................................................. 13 CHAPTER 3 ........................................................................................................................................ 16 3. Antenna design .................................................................................................................................. 16 3.1 Design Specifications ................................................................................................................... 16 3.2 Design of half wavelength dipole antenna .................................................................................... 17 3.3 Parametric study of simple dipole antenna .................................................................................. 17 3.3.1 Effect of changing feeder length ‘fL’ ....................................................................................... 17 3.3.2 Effect of changing length of antenna ‘L’................................................................................. 18 3.3.3 Effect of changing width of antenna ‘W’ ................................................................................ 19 3.3.4 Effect of changing feed gap ‘fgap’ ............................................................................................ 20 3.4 Design of bowtie dipole antenna .................................................................................................. 20 3.4.1 Parasitic element .................................................................................................................. 21 3.5 Design approach and simulation results ....................................................................................... 22 3. 6 parametric study of bowtie dipole antenna without parasitic element ........................................ 22 3.6.1 Effect of changing length of antenna ‘L’................................................................................. 22 3.6.2 Effect of changing width of antenna ‘W’ ................................................................................ 23 3.6.3 Effect of changing feeder length ‘fL’ ....................................................................................... 24.

(10) vi 3.6.4 Effect of changing feed gap ‘fgap’ ............................................................................................ 25 3.7 Comparison of simulated results of bowtie dipole antenna with and without parasitic element ... 25 3.8 Parametric study of bowtie dipole antenna with parasitic element .............................................. 26 3.8.1 Effect of changing length of the parasitic element ‘Lpar’ ......................................................... 26 3.8.2 Effect of changing width of the parasitic element ‘Wpar’ ........................................................ 27 CHAPTER 4 ........................................................................................................................................ 28 4. Differential coupler design ................................................................................................................ 28 4.1 Hybrid coupler design ................................................................................................................. 28 4.2 90° Hybrid coupler ....................................................................................................................... 28 4.3 180° Hybrid coupler ..................................................................................................................... 30 4.4 180° differential coupler .............................................................................................................. 32 4.5 Wilkinson power divider design ................................................................................................... 34 4.5.1 Wilkinson low band power divider (1.6 GHz) ......................................................................... 34 4.5.2 Wilkinson high band power divider (2.6 GHz) ........................................................................ 36 CHAPTER 5 ........................................................................................................................................ 39 5. Integrated design ............................................................................................................................... 39 5.1 Introduction ................................................................................................................................. 39 5.2 Polarization.................................................................................................................................. 39 5.3 Isolation versus polarization......................................................................................................... 40 5.4 Isolation versus distance .............................................................................................................. 42 5.5 Integration of fully differential MIMO antenna with and without fully differential 180o coupler ... 43 CHAPTER 6 ........................................................................................................................................ 46 6. Results and discussion ....................................................................................................................... 46 6. 1 Results and discussion on modified bowtie dipole antenna design .............................................. 46 6.2 Simulated results ......................................................................................................................... 47 6.3 Prototype of differential MIMO antenna ...................................................................................... 48 6.4 Integrated prototype ................................................................................................................... 49.

(11) vii CHAPTER 7 ........................................................................................................................................ 51 7. Conclusion and future work ............................................................................................................... 51 7.1 Conclusion ................................................................................................................................... 51 7.2. Future work ................................................................................................................................ 51.

(12) 1. ____________________________________ CHAPTER 1. 1. Introduction Wireless technologies are growing rapidly day by day in today’s communication era [1]. During the last decades the innovations of new technologies have been escalating. In the present generation the consumer demands are increasing for better compatibility of mobile devices which consumes less power, faster data transfer rate, better features and still compact in size so that it will be convenient to carry and use. With the advent increase in consumer demands day by day there is a need to employ a technology that provides a high data rate. By using multiple antennas that instead of one antenna at mobile terminal yields higher throughput and communication performance [2]. Therefore MIMO technology, the use of multiple antennas at both the transmitter and receiver end has attracted attention much in wireless communication. Because of these properties, MIMO technology has also been found to be important in the wireless communication standards such as IEEE 802.11n, 4G, 3GPP LTE and WiMAX [3]. Due to the possibility of MIMO operation good isolation and correlation between antennas are required. Since in MIMO technology multiple antennas are used at transmitter and receiver, in the thesis (i.e. two bowtie dipole antennas) is considered and also good isolation between antennas is required. So that each antenna receives an independent signal and there by the antennas perform at their maximum capability. It is known that the correlation between antennas has direct impact on distance between antennas i.e., the more the distance between antennas the greater isolation. Employing MIMO technology in wireless portable devices is limited to size. Therefore fully differential 180 o coupler was introduced to neutralize the coupling between antennas therefore isolation is improved.. 1.1 Background This project is done for the fulfillment of Master’s degree in Electrical Engineering at Linköping University. As the users of the wireless handheld devices are growing rapidly during the last decade there is a demand for high data rate and smaller size of the communication devices. It is possible to increase the overall system performance by employing the MIMO antenna technology inside a hand held device, i.e., higher throughput and flexibility. In order to increase the isolation between the antennas in wireless portable devices a decoupling network is introduced which reduces the coupling between the antennas which in turn improves the isolation between the antennas..

(13) 2. 1.2 Objective o. The main objective of this thesis is to integrate a MIMO antenna with fully differential 180 hybrid coupler in the frequency range of 0.7-3 GHz by using the Electronic Design Automation (EDA) and Computer Aided Design (CAD) tool ADS 2009 from Agilent Technologies Inc. In the process a MIMO antenna (i.e. two bowtie dipole antennas) are designed in the frequency range of 0.7-3 GHz and simulated to see the performances of the antenna in terms of input reflection, o radiation pattern and efficiency. In the next part differential 180 hybrid coupler in the frequency range of 0.7-3 GHz is designed and simulated to see the performance of input reflection, phase o difference and isolation. The last part is to integrate MIMO antenna with a fully differential 180 hybrid coupler and to compare the results of isolation between antennas with and without coupler.. 1.3 Thesis overview It is comprised of seven chapters each chapter presents in detail information on different aspect related to the thesis. Chapter 1: It presents introduction in basically an overview of the thesis followed by background, objective which is the most important for what is to be achieved and the last part is the thesis overview which is information regarding the outline of the thesis.. Chapters 2: It explains in brief about antenna basic parameters in order to get the theoretical knowledge regarding the antenna which would be necessary in the design process. It also explains the brief theory of basic 90° hybrid coupler, 180° hybrid coupler and 180° differential coupler. Chapter 3: It focuses on the design procedure and simulated results of simple dipole antenna and bowtie antenna followed by parametric study. Chapter 4: It focuses on the design procedure of basic 90° hybrid coupler, 180° hybrid coupler, 180° differential hybrid coupler design based upon the requirements and also the design of Wilkinson power dividers on low and high band frequencies. Chapter 5: It is integrated design of dipole antenna with 180° differential coupler and the simulation results are explained with and without coupler. Chapter 6: This chapter is about the results and discussions of bowtie antenna, 180 differential coupler and integrated design.. o. Chapter 7: Conclusion of the thesis followed by future work recommendation and references..

(14) 3. ____________________________________ CHAPTER 2. 2. Theory 2.1 Antenna Antenna’s plays a major role in communications for both receiving and transmitting of signals. It is a transducer which converts guided electromagnetic wave in wave guide or transmission line to an electromagnetic wave propagating in free space or vice versa. Antennas exhibit reciprocity property which means the receiving and transmitting radiation characteristics are the same. The antennas are designed depending upon their applications for wide band or narrowband while considering the different types of properties such as input reflection, directivity, and radiation pattern and antenna gain. The different key parameters related to antennas are explained in brief below.. 2.1.1 Input impedance Antenna input impedance is defined as “the impedance presented by an antenna at it terminals or else the ratio of voltage to current” [5]. The characteristic impedance should be same for all the components connected with antenna and the transmission cable in order to occur maximum power transfer [4]. In general transmitters are designed typically 50 Ω impedance for efficient and safe operation of the circuitry. Antenna mismatch occurs if the antenna input impedance is not matched with the source, e.g., 50, 75 Ω etc.. 2.1.2 Bandwidth It is defined as the range of frequencies in which antenna can radiate or receive energy. It is determined by measuring standing wave ratio (SWR) where it is less than 2:1. The SWR over 2:1 indicates that the antenna is no longer performing efficiently, so keeping SWR under 2:1 is considered in antennas bandwidth. Bandwidth can also be considered as range of frequencies on either side of center frequency where the antenna characteristics such as input impedance, gain, radiation efficiency, polarization pureness are commonly evaluated parameters [5]. Bandwidth can also be displayed as relative bandwidth in terms of percentage as follows Eq. (2.1) Where fh = Highest frequency fL = Lowest frequency.

(15) 4. fc = Centre frequency which is given by Eq. (2.2). 2.1.3 Input reflection The physical meaning of S11 (dB) is the input reflection coefficient with output of network terminated by matched load. Input reflection is measured using S-parameters.. Eq. (2.3) Where SWR = Standing wave ratio.. 2.1.4 VSWR Voltage Standing wave ratio is defined as ratio of maximum voltage (current) to minimum voltage (current) and is unit less quantity. Eq. (2.4) VSWR is a measure of how well the antenna is matched to transmission line it connects to. A low VSWR means the antenna is well matched and more power is delivered to the antenna. In addition to that high VSWR means that the signal is reflected back to the source. A VSWR value of 1 means all the power reaching destination with no reflection, which is the ideal case. But in the real world VSWR value of 2 is acceptable.. 2.1.5 Directivity Directivity is defined as “the ratio of radiation intensity in a given direction from antenna to that of average radiation intensity in overall direction”. The average radiation intensity of an antenna is equal to power radiated by an antenna divided by 4π [5], [7]. Directivity is expressed as Eq. (2.5) Where D = Directivity Umax = Maximum Radiation Intensity Prad = Radiated Power. 2.1.6 Gain Antenna gain is the important parameter that determines how much it is radiated in one particular direction. It is defined as “the ratio of the radiation intensity in given direction to that to the radiation intensity that would be obtained if the power accepted by the antenna were radiated isotropically”.

(16) 5. [5]. The radiation intensity corresponding to isotropically radiated power is given by input power (Pin) divided by 4π. Antenna gain is expressed as Eq. (2.6) Eq. (2.7) Where G = Gain of antenna Pin = Input power D = Direction of antenna ec = Conduction efficiency ed = Dielectric efficiency U = Maximum Radiation Intensity Therefore the antenna is designed depending upon the application of antenna. For the design of high gain antenna the quality of signal and the direction would be the main criteria whereas for low gain antenna the orientation of antenna is unimportant.. 2.1.7 Radiation intensity Radiation intensity is defined as “the power radiated from an antenna per unit solid angle” [3]. It is expressed as Eq. (2.8) Where U = Intensity of Radiation (Ws/r) Wrad = Radiation density (W/m2). 2.1.8 Front to back ratio It is the ratio of signal strength transmitted in forward direction to that of signal strength transmitted in backward direction. “The front to back end ratio is the difference in dB between the level of the maximum radiation and the level of radiation in a direction 1800” [9].. 2.1.9 Radiation pattern The radiation pattern of antenna at fixed or constant distance illustrates the relative strength of the radiated field from the antenna in various directions. It is represented in two dimensional or three dimensional and the pattern measurements are shown in rectangular or polar format. As Isotropic.

(17) 6. antenna radiates equally in all directions then there radiation pattern will be like a sphere and in case of dipole or monopole the power will be emitted in horizontal directions. Figure 2.1 shows the radiation pattern of the antenna. The radiation pattern for different types of antenna will be like lobes at various angles separated by nulls as shown in Figure 2.1 and the properties of radiation includes field strength, radiation intensity, power flux density, phase and polarization [5].. Figure 2.1. Radiation pattern of antenna [6]. 2.1.10 Side lobes The antenna radiates all the energy in not only one particular direction it radiates some energy in other directions away from the main lobe those peaks are called as side lobes. The power density in main lobes will be higher than that of side lobes. Side lobes are commonly specified in dB.. 2.1.11 Polarization The polarization of antenna is defined as the orientation of an electric field of an electromagnetic wave with respect to earth surface [6]. Polarization is effected due to reflections. If the polarization of the transmitting and receiving antenna is same then the maximum power transfer takes place between transmitting and receiving antenna. The overall efficiency and the performance will be reduced when there is reduction in power transfer which takes place when the transmitting antenna and receiving antenna polarizations are different.. 2.1.12 Beamwidth The half power beam width is the angle across the main lobe of an antenna pattern between two directions at which radiation intensity is half of it‟s the maximum value at the center of lobe. Half power beam width is expressed in decibels dB [4].. 2.1.13 Field regions The space adjoining the antenna is usually divided in to three regions reactive near-field, the radiating near-field and the far-field regions. These regions have their individual characteristics and.

(18) 7. designed to make out the field structure in each region but no abrupt changes are noted in field configurations.. 2.1.13.1 Reactive near-field “The portion of near field immediately surrounds the antenna where the reactive field dominates” [5]. In this region energy decays rapidly with the distance.. 2.1.13.2 Radiating near-field “The region of the near field of antenna between reactive near field region and far field region where the radiation field is predominate and angular field distribution is dependent upon the distance from the antenna” [5].. 2.1.13.3 Far field region “The region of the field of the antenna where angular field distribution is independent of the distance from the antenna” [5]. If the maximum dimension is D which is large compared to the wavelength then the far field region is expressed as Eq. (2.9) Where D = Largest dimension of antenna λ = Wavelength. 2.1.14 Antenna efficiency It is defined as the ratio between powers radiated from antenna to the power at input terminals [5]. Antenna efficiency is expressed as follows Eq. (2.10) Where eo = Total efficiency er = Reflection efficiency ec = Conduction efficiency ed = Dielectric efficiency. 2.1.15 Envelope correlation The measure of this independency between antennas is termed as correlation which is denoted by, the value of lies between 0 and 1. Generally correlation less than 0.5 are preferred in MIMO systems, because to install more antennas within a small space and still keep them working when the channel.

(19) 8. scattering is rich enough. So the correlation coefficient between any pair of antenna elements should be less than 0.5. The calculation of envelope correlation between antennas can be approached in different ways [26]. One is based on far filed pattern and the other is by using S-parameters. The third one is based on Clarke’s formula. Correlation using far field pattern is time consuming since we require measurement of radiation pattern. Therefore simple method of calculating correlation value is by using S-parameters and is given by [27].. Eq. (2.11) Where S11 and S22 are respective input reflections of antennas S21 is isolation between antennas.. 2.2 Dipole antenna Dipole antenna is a balanced antenna which is made up of two straight wires which are on same axis which are equal in length and extended in opposite directions with a small gap in between them and is called feed gap [11]. The dipole antenna is mostly fed at center known as feeder through which power is applied for transmitting and receiving of electromagnetic radiation [12]. Figure 2.2 shows half wavelength dipole antenna structure which is made of two equal wires with equal length has an electrical length of half a wavelength as results this is called as half wave dipole antenna.. Figure 2.2. Half wavelength dipole antenna. The lengths of the both wires are equal so the total length of the dipole antenna is half wavelength which divides each section of dipole to quarter wavelength long [13].. 2.2.1 Antenna length Antenna length is the main factor for determining the operating frequency of an antenna. The length for a wave travelling in free space for a half wave dipole is calculated and multiplied by a factor A..

(20) 9. Figure 2.3 show the graph for determining factor A. Since the micro strip dipole antenna is thin when compared to wavelength, from the Figure 2.3 wavelength (λ) to thickness (d) yields a high value of 0.98 [12].. Figure 2.3. Graph for determining factor A. The length of half wave dipole (Lλ/2) can be calculated by the formula Eq. (2.12) Depending upon the thickness of the wire the length of the antenna changes, it is better to make slightly long because we can later trim it to operate at desired frequency.. 2.2.2 Dipole feed impedance The most important factor dipole feed impedance is at the point in antenna where the feeder is connected. Feed impedance for dipole antenna is determined where the voltage is maximum and current is minimum in half wave dipole antenna. In between feeder and antenna maximum power is transferred only when the antenna and feed impedance are matched. The feed impedances of dipole antenna can be varied depending upon the various factors such as feed position, ground and length. The feed impedance is measured as 73.13 Ω in a half wave center fed dipole antenna making it ideal to feed with 75 Ω feeder [12].. 2.2.3 Wideband dipoles In order to operate the antenna over the desired frequency range, a dipole antenna with wider band characteristics are required. There are many ways of designing dipole antennas with wideband characteristics namely dipole array antennas, spiral dipole antennas, biconical antennas. Biconical antenna would be preferable one if considering the requirement of small structure. However, the design of biconical antenna is impractical due to fact that shell structure is massive. Instead of it a bowtie antenna which is an approximation of biconical one is found to be an interesting one [13]..

(21) 10. 2.2.4 Parasitic element A passive radiator or parasitic element is a conductive element which is not electrically connected to anything else i.e., does not have any wired input. The purpose of parasitic element is to improve radiation pattern of an antenna and to enhance the impedance bandwidth. The parasitic element absorbs the radio waves from the active element and re-radiates i.e., the main radiator couples to the parasite element, e.g., capacitive coupling. This changes the radiation pattern of antenna and the gain and performance of antenna increases [14]. Inappropriate placement or else the geometry of parasitic element would cause degradation in overall antenna’s performance. So the parametric study of parasitic element would be the best choice [15].. 2.3 Hybrid coupler Hybrid couplers are widely used in radio frequency circuit design for power combing and power divisions. It is categorized into three port component and four port components. The main advantage of using a hybrid junction is because of its power handling capability. A 90° hybrid coupler is a four port component which functions as power divider or combiner with 90° phase difference and used in applications such as balanced mixers and antenna array fed networks [28]. The 180° hybrid coupler is also called as rat race coupler which is widely used in design of balanced mixers. As the main focus is on the 180° differential coupler where there is an increase in bandwidth when compared to single ended coupler. Differential coupler is difficult to understand because of its complex size and structure which involves both broadside coupling and edge coupling. In order to achieve tight coupling it uses a combination of edge coupling involving adjacent strip conductors and broadside coupling between strip conductors on different metallization layers [29].. 2.3.1 90° Hybrid coupler It consists of four ports, port 1 is input port, port 2, 3 are output ports and port 4 is isolation port. In which port 2 and 3 has equal power division and phase difference of 90° between the ports. When all ports are matched power enters from port 1 and delivers in port 2 and 3 equally with 90° phase difference between the ports and the port 4 in which there will be no power is coupled which is isolation port [30]. Figure 2.4 shows the 90° hybrid coupler in which any port can be used as input port due to the design symmetry when one port acts as input port the other opposite port from left to right, will be the output ports and the remaining port is isolated port on the same side as input port. When port 1 is input port this enters and when mismatch occurs at port 2 the power is reflected back and divides in port 1 and 3 in this the port 4 acts as isolation port..

(22) 11. Figure 2.4. 90° Hybrid coupler. Figure 2.5 shows the schematic of quadrature 90° hybrid coupler is shown below in which each line symbolize transmission line with characteristic impedance normalized to Zo.. Figure 2.5. Circuit of the branch line hybrid coupler in normalized form. The amplitudes of the incident waves for these ports can be given as B1 = ½ Гe + ½ Гo. Eq. (2.13). B2 = ½ Te - ½ To. Eq. (2.14). B3 = ½ Te - ½ To. Eq. (2.15). B4 = ½ Гe - ½ Гo. Eq. (2.16). Where Гe,o and Te,o are even mode and odd mode reflection and transmission coefficients for the two port network [30]. By calculating the even mode reflection coefficient and odd mode reflection coefficient of even mode two port circuit which can be done by multiplying each cascade component in that circuit..

(23) 12. =. Eq. (2.17). Reflection and transmission coefficients can be obtained by using formula Гe =. =. Te =. =. =0. Eq.(2.18). =. Eq. (2.19). Similarly for odd transmission and reflection coefficient are given as follows =. Eq. (2.20). Then we get the reflection and transmission coefficients as Гe = 0. Eq. (2.21). Te =. Eq. (2.22). Therefore we get by using the formulas in the above equations B1 = 0. Eq. (2.23). B2 =. Eq. (2.24). B3 =. Eq. (2.25). B4 = 0. Eq. (2.26). So first row of S parameter matrix can be written as [S1] = [0. 0]. Eq. (2.27). The above values of B1, B2, B3, B4 agrees with first row and the other row of scattering parameter matrix can be obtained by transposition of first row due to high degree of symmetry. Thus the Scattering [S] matrix for the quadrature hybrid coupler is given as. S. 0 j 1 0 1 j 0 0 1 2 1 0 0 j 0 1 j 0. Eq. (2.28).

(24) 13. 2.3.2 180° Hybrid coupler The 180° hybrid coupler is also called as rat race coupler which is a four port device depending up on which port is excited it has either in phase 0° or out of phase 180° phase difference between the output ports. Figure 2.6 shows the symbol of 180° hybrid coupler when signal is applied at port 1 it equally splits in phase components at port 2, 3 and the port 4 remains as isolated port.. Figure 2.6. Symbol for 180° hybrid junction. Similarly when the input signal is applied at port 4 it equally split with 180° phase difference between port 2 and 3 and port 1 is isolated port [31]. Figure 2.7 shows ratrace coupler which consists of three λ/4 sections and one λ/4 section and the port impedances are of 50 Ω and the ring characteristic impedance is of 50√2 Ω.. Figure 2.7. 180° Ratrace coupler. The amplitudes of the scattered waves in ring hybrid structure will be given as B1 = ½ Гe + ½ Гo. Eq. (2.29). B2 = ½ Te +½ To. Eq. (2.30). B3 = ½ Г e - ½ Г o. Eq. (2.31). B4 = ½ T e - ½ T o. Eq. (2.32). Then we can obtain reflection and transmission coefficients using ABCD matrix for odd and even mode two port circuits gives the results shown below.

(25) 14. Eq. (2.33). Eq. (2.34) We get the results as Гe =. Eq. (2.35). Гe =. Eq. (2.36). Гo =. Eq. (2.37). Гo =. Eq. (2.38). Therefore we get these results by using the formulas in the equations B1 = 0. Eq. (2.39). B2 =. Eq. (2.40). B3 =. Eq. (2.41). B4 = 0. Eq. (2.42). From the above indicates input port is matched and port 4 is isolated thus input power is divided in phase between ports 2 and 3. Thus we get the results for the first row and colum of scattering or Sparameters. By considering unit amplitude wave incident at port 4 of ring hybrid. Then the scatterd waves will be B1 = ½ Гe - ½ Гo. Eq. (2.43). B2 = ½ Гe - ½ Гo. Eq. (2.44). B3 = ½ Гe + ½ Гo. Eq. (2.45). B4 = ½ Гe +½ Гo. Eq. (2.46). The ABCD matrix for even and odd mode circuits Eq. (2.47). Eq. (2.48).

(26) 15. Thus reflection and transmission coefficents are Гe =. Eq. (2.49). Гe =. Eq. (2.50). Гo =. Eq. (2.51). Гo =. Eq. (2.52). Therefore we get these results by using the formulas in the equations B1 = 0. Eq. (2.53). B2 =. Eq. (2.54). B3 =. Eq. (2.55). B4 = 0. Eq. (2.56). From the above indicates input port is matched and port 1 is isolated thus input power is divided in phase between ports 2 and 3 with 180o phase difference. Thus we get the results for the fourth row and column of Scattering martix. By usng symmetry consideration the remaining elements can be found for scattering matrix. The Sacattering martrix for 3 dB 180o hybrid coupler is given as. 0 1 1 0 S    1  2 1 0  0  1. 1 0 0  1 0 1  1 0. Eq. (2.57).

(27) 16. ____________________________________ CHAPTER 3. 3. Antenna design 3.1 Design Specifications The RO 4360 substrate is selected from the different available substrates in the ITN PCB laboratory of Linköping University for both design of dipole antenna and differential coupler. The higher dielectric constant of the substrate is chosen which reduces the size of the circuit as the thesis is to integrate antenna with coupler in compact dimension. Measurements were done by using an Rhode and Schwartz ZVM vector network analyzer. Table 3.1 : Substrate definition Parameter (Rogers 4360). Dimension. Dielectric constant. 6.15. Substrate thickness. 0.305 mm. Copper thickness. 18.0 µm. Loss tangent. 0.003. Copper conductivity. 5.8 x 10^7 S/m. Conductor surface roughness. 1.0 µm. Table 3.2 : MIMO Antenna requirements Specification. Range. Frequency band. 0.7- 3 GHz. Input reflection. < -10.0 dB. VSWR. < 2.0. Correlation. < 0.5. Isolation. < -20.0 dB.

(28) 17. 3.2 Design of half wavelength dipole antenna Figure 3.1 shows the layout of simple half wavelength dipole antenna which consists of two legs which are equal in size with ‘L/2’ being the length of each arm of the dipole antenna and width is defined as ‘W’.. Figure 3.1. Layout of half wavelength dipole antenna. Where, L/2 = Length of each arm of the dipole antenna W = width of dipole antenna fgap = feed gap fL= feeder length. 3.3 Parametric study of simple dipole antenna Varying one parameter at a time, keeping all the other parameters fixed, and observing how the change in one single parameter effects the antenna performance is called parametric study, i.e., to isolate the dependence of each variable. First the parametric study of simple dipole antenna is conducted. By varying one parameter say the length and all the other parameters fixed and observed in the schematic how the radiation pattern is changed by means of varying single parameter [20].. 3.3.1 Effect of changing feeder length ‘fL’ In this study only feeder length is varied and the values of the other parameters such as L/2 = 35 mm, W = 10 mm, fgap = 0.57 mm are fixed. The line calculator in ADS it was found that if the substrate parameter is defined and operating frequency is kept 1.85 GHz the feeder length of 2.0 mm is obtained. Then feeder length is varied in three different cases, while the other parameters remains fixed and the simulated results are observed with respect to input reflection (dB)..

(29) 18. fixed and the simulated results are observed with respect to input reflection (dB). Figure 3.2 illustrates the simulated results for variation of input reflection over three different values of fL. From the figure 3.2 it is observed that when the feeder length changes from 4.7 to 2.0 mm there was shift in the resonant frequency and input reflection. When it is varied from 2.0 to 3.0 mm there was much change in the input reflection and slight shift in the resonant frequency.. Figure 3.2. Simulated results for variation of input reflection over feeder length ‘fL’. 3.3.2 Effect of changing length of antenna ‘L/2’ In this study the antenna length (L/2) is varied and the values of the other parameters such as f L = 2.0 mm, W = 10.0 mm, fgap = 0.57 mm are fixed. A half wavelength antenna for a substrate with εr is given by. Where εr was used as εeff to get the initial design value for the dipole antenna arm length, and f c is center frequency of 1.85 GHz. The calculated value of L/2 by using the above equation is found to be 35 mm. Figure 3.3 illustrates the simulated results for variation of input reflection over three different values of half wavelength antenna (L/2). The corresponding value of input reflection in three various cases are shown in graph. Firstly the antenna is simulated by using the half wavelength L/2 = 33.0 mm where it can see from the simulated results that center frequency is at 1.926 GHz. Then after the length is changed to be 30.0 mm it is noticed that there is shift in the center frequency to 2.234 GHz. This variation is because as the length of antenna increases the frequency decreases and vice versa. After that the half wavelength of antenna (L/2) is increased from 30.0 to 35.0 mm where the center frequency is shifted to 1.85 GHz which is the desired one..

(30) 19. Figure 3.3. Simulated results for variation of input reflection over half wavelength antenna (L/2). 3.3.3 Effect of changing width of antenna ‘W’ In this study the antenna width (W) is varied and the values of the other parameters such as fL = 2.0 mm, L/2 = 35.0 mm, fgap = 0.57 mm are fixed. Figure 3.4 illustrates the simulated results for variation of input reflection over three different values of antenna width (W). Antenna width is varied in three different cases it is noticed that when width of antenna increases the bandwidth of the antenna increases and the value of input reflection also increases which means the performance of antenna improves. When the width changes from 15.0 mm to 13.0 mm the bandwidth of antenna has decreased and even the input reflection value has decreased. When it is further decreased to 10.0 mm this results in degradation of antenna performance.. Figure 3.4. Simulated results for variation of input reflection over antenna width ‘W’.

(31) 20. 3.3.4 Effect of changing feed gap ‘fgap’ In this study the feed gap (fgap) is varied and values of the other parameters such as fL = 2.0 mm, L/2= 35.0 mm, W = 15.0 mm are fixed. Figure 3.5 illustrates the simulated results for variation of input reflection over three different values of feed gap (fgap). As the gap between feeder arms of dipole antenna increases the bandwidth increases and the input reflection decreases.. Figure 3.5. Simulated results for variation of input reflection over ‘fgap’. 3.4 Design of bowtie dipole antenna The concept of bowtie antenna is first introduced by Oliver lodge in 1898. Over the many years work has been done to transform simple dipole to obtain a wideband characteristics and such work includes design of various dipole antennas such as bowtie antenna, butterfly and diamond shape dipole antenna. But the bowtie antenna is most extensively used because of the ease of geometry and high radiation efficiency. Bowtie antenna design was popularized by Brown and Woodward in 1950s [17]. Figure 3.6 shows basic design of bowtie dipole antenna. It consist of two triangular shaped elements which are fabricated on a same single substrate whereas in micro strip antenna consist of two rectangular patches. The advantage of bowtie antenna when compared to normal wire dipole antenna is it inherently possesses wider bandwidth and can be used for dual bad applications..

(32) 21. Figure 3.6. Design of basic bowtie dipole antenna. The half wavelength of bowtie dipole antenna for a substrate with εr is given by. Where εr was used as εeff to get the initial design value for the bowtie dipole antenna arm length and fc is center frequency which is calculated by using the formula Eq. (3.3) Where fh is the highest frequency = 3 GHz and fL is the lower frequency = 0.7 GHz Therefore fc is calculated to 1.85 GHz.. Bowtie antennas are used for many applications such television, Ground Penetrating Radar (GPR) and dual band applications [18]. Bowtie antennas have many advantages such as low profile, high radiation efficiency, low manufacturing cost and ease of fabrication [19].. 3.4.1 Parasitic element The design of a parasitic element generally utilizes the method of arranging the antenna and a parasitic element together on the same dielectric material. Initially the parasitic element is separately designed and optimized by carefully placing it at conventional place in order to achieve the desired bandwidth [23], [24]. This approach is more advantageous since addition parasitic element changes the radiation pattern of antenna which results in an improvement in overall antenna performance without degrading the overall system characteristics. Inappropriate placement and geometry of the parasitic element would cause changes in the resonance characteristics of existing internal antenna and these results in degradation of the antenna’s performance. So the parametric study of parasitic element would be the better approach to enhance the performance of the antenna..

(33) 22. 3.5 Design approach and simulation results Figure 3.7 shows the geometry of the designed bowtie dipole antenna with parasitic element. The bowtie antenna was designed on R04360B substrate with a relative dielectric constant of εr = 6.15 and a thickness (h) = 0.305 mm. The two arms of bowtie antenna have equal length and width and parasitic element of Length Lpar and width Wpar.. Figure 3.7. Geometry of the designed bowtie dipole antenna with parasitic element. L/2 = Length of each arm of the bowtie dipole antenna W = Height of bowtie dipole antenna fw = Feeder width fL = Feeder length fgap = Feed gap Lpar = Length of parasitic element Wpar = Width of parasitic element. 3. 6 parametric study of bowtie dipole antenna without parasitic element 3.6.1 Effect of changing length of antenna ‘L/2’ In this study the antenna length (L/2) is varied and the values of the other parameters such W = 10.0 mm, fL = 4.0 mm, fW = 0.6 mm, fgap = 0.37 mm are fixed. Figure 3.8 illustrates the simulated results for variation of input reflection over three different values of antenna length (L/2). The calculated value of the antenna length is found to be L/2 = 35.0 mm. The length of antenna is varied ín three different cases. Firstly the antenna is simulated by using length (L/2) of 35.0 mm and it is.

(34) 23. noticed from the simulated results that center frequency is at 1.926 GHz. Then after the length is changed to be 50.0 mm where it is noticed that there is shift in the center frequency to 1.231 GHz. Finally the length (L/2) was increased to 63.0 mm where the frequency is shifted to 959 MHz .The final conclusion is found was as the length of the antenna increases the frequency shifts to lower frequency band. This variation is the frequency shift is because the length and frequency are cross related to each as the length increases frequency decreases and vice versa.. Figure 3.8. Simulated results for variation of input reflection over antenna length ‘L/2’. 3.6.2 Effect of changing width of antenna ‘W’ In this study the antenna width (W) is varied and the values of the other parameters such are L/2 = 63.0 mm, fL = 4.0 mm, fW = 0.6 mm, fgap = 0.37 mm are fixed. Figure 3.9 illustrates the simulated results for variation of input reflection over three different values of antenna width (W). From the simulated results when the width of antenna increased from 15.0 mm to 18.75 mm there was increase in bandwidth and input reflection. However further increase in width results in decrease of bandwidth as observed in above simulated results. The optimized value of W = 18.75 mm..

(35) 24. Figure 3.9. Simulated results for variation of Input reflection over antenna width ‘W’. 3.6.3 Effect of changing feeder length ‘fL’ In this study only feeder length is varied and the values of the other parameters such L/2 = 63.0 mm, W = 18.75 mm, fW = 0.6 mm, fgap = 0.37 mm are fixed. The line calculator in ADS it was found that if the substrate parameter is defined and operating frequency is kept 1.85 GHz the feeder length of 2.0 mm is obtained. Then feeder length is varied in three different cases, while the other parameters remains fixed and the simulated results are observed with respect to input reflection (dB). Figure 3.10 illustrates the simulated results for variation of input reflection over three different values of fL. From the simulated result it is observed that when feeder length is changed from 2.0 to 4.0 mm the resonant frequency has been shifted to lower frequencies and when it is further increased to 8.0 mm the resonant has been shifted still towards lower frequencies and there was much increase in bandwidth and input reflection when compared to the other two cases.. Figure 3.10. Simulated results for variation of return loss over antenna feeder length‘fL’.

(36) 25. 3.6.4 Effect of changing feed gap ‘fgap’ In this study the feed gap (fgap) is varied and the values of the other parameters such as L/2 = 63.0 mm, W = 18.75 mm, fW = 0.6 mm, fL = 8.0 mm are fixed. Figure 3.5 illustrates the simulated results for variation of input reflection over three different values of feed gap (fgap). As the gap between feeder arms of dipole antenna increases the bandwidth increases and the input reflection decreases. As the feed gap increases from 0.31 to 0.37 mm there was an increase in impedance bandwidth, i.e., input reflection. However further increased in feed gap from 0.37 to 0.43 mm results in decrease in input reflection and impedance bandwidth. So the optimized value of feed gap based on the study is found to be 0.37 mm.. Figure 3.11. Simulated results for variation of input reflection over antenna feed gap ‘fgap’. 3.7 Comparison of simulated results of bowtie dip antenna with and without parasitic element Figure 3.12 shows simulated results of the bowtie dipole antenna with and without parasitic element. It is observed that the antenna without parasitic element is not operating under the desired frequency band 0.7 - 3 GHz, since the input reflection value is much higher over the desired frequency band but when a parasitic element is added to the antenna the performance of the antenna is much improved (increase in bandwidth almost 1 GHz) this can be clearly observed in Figure 3.12..

(37) 26. Figure 3.12. Simulated results of bowtie dipole antenna with and without parasitic element. It is observed that the bowtie dipole antenna with parasitic element has wider impedance bandwidth when compared to antenna without parasitic element. Addition of parasitic element increases the bandwidth as long as the length and width property are chosen.. 3.8 Parametric study of bowtie dipole antenna with parasitic element 3.8.1 Effect of changing length of the parasitic element ‘Lpar’ In this study the length of parasitic element was varied in three different cases while all the other parameters remained unchanged and their respective values are L/2 = 63.0 mm, W = 18.75 mm, f L = 8.0 mm, fW = 0.6 mm, fgap = 0.37 mm and Wpar = 12.0 mm. Figure 3.13 shows the simulated results for variation of input reflection over the different lengths of parasitic element. It can be observed that the resonance frequency remains unchanged in all the three different cases, while there is much shift in the input reflection and bandwidth depending on the parasitic length. When the length of the parasitic element was increased from 32.0 to 38.5 mm there was much improvement in input reflection in the desired bandwidth. Further increasing the length of the parasitic element from 38.5 to 44.0 mm still results in much more improvement in input reflection and bandwidth. So the optimized value of length of parasitic element is found to be 44.0 mm and if further increased in length the performance remains unchanged..

(38) 27. Figure 3.13. Simulated results for variation of input reflection over length of parasitic element. 3.8.2 Effect of changing width of the parasitic element ‘Wpar’ In this study the width of parasitic element was varied in three different cases while all the other parameters remained unchanged and their respective values are L/2 = 63.0 mm, W = 18.75 mm, f L = 8.0 mm, fW = 0.6 mm, fgap = 0.37 mm and Lpar = 44.0 mm. Figure 3.14 shows the simulated results for variation of input reflection over the three differentvalues of the width of the parasitic element. It is found that when Wpar is increased from 12.0 to 15.0 mm there was much increase in return loss while the bandwidth was not affected. When it is further increased to 17.5 mm both the input reflection and the bandwidth is improved. After the parametric study of width of bowtie dipole antenna the optimized value is found to be 17.5 mm. Therefore the finalized antenna parameters after parametric study are L/2 = 63.0 mm, W = 18.75 mm, fL = 8.0 mm, fW = 0.6 mm, fgap = 0.37 mm, Lpar = 44.0 mm, Wpar = 17.5 mm .. Figure 3.14. Simulated results for variation of input reflection over the width of the parasitic.

(39) 28. ____________________________________ CHAPTER 4. 4. Differential coupler design 4.1 Hybrid coupler design One of the main parts of the thesis was to design a 180° differential coupler which could operate in the 0.7-3 GHz frequency band. In this thesis to present a fully differential two antenna MIMO with good isolation between antennas. Generally when the distance between the MIMO antennas increases the isolation increases but in handheld devices the size of circuitry is restricted to a relative small size. In order to reduce the circuitry the antenna are kept close to each other so that the isolation decreases in order to have good isolation the decoupling network is introduced in between antennas. The role of the 180° differential coupler is to neutralize the coupling between the antennas as isolation is increased. In this section before design of 180° differential coupler the 90° hybrid coupler, 180° hybrid coupler were designed and simulated as to understand the principle behind them after analyzing both hybrid couplers 180° differential coupler is designed and simulated according to the requirements.. 4.2 90° Hybrid coupler The 900 hybrid coupler is designed in the frequency range of 0.7-3 GHz frequency band. In accordance to the theory the four transmission lines of 90° hybrid coupler are quarter wavelength where two of the having characteristic impedance of Zo/√2 and Zo where Zo consider to be 50 Ω. The length and widths of transmission line is calculated using line calculator at an operating frequency of 1.85 GHz. Figure 4.1 and 4.2 shows the schematic and layout for the 90° hybrid coupler. The line calculator in ADS it was found that if the substrate parameter is defined and operating frequency is kept 1.85 GHz the length and width to transmission line are obtained. The length and width of TL1 and TL2 are 18.9 mm and 0.79 mm .The length and width of TL1 and TL2 are 19.12 mm and 0.45mm..

(40) 29. o. Figure 4.1. Schematic of 90 hybrid coupler. Figure 4.2. Layout of 90° hybrid coupler. Figure 4.3 shows simulation results of single ended 90° coupler design in schematic level. In figure 4.3 (a) shows the S-parameters S21, S31 there is an equal 3 dB split at centre frequency, S 11, S41 with the isolation of -34.18 dB and isolation S41 of -40 dB at center frequency. The phase difference in of o 90 is shown in figure 4.3(b)..

(41) 30. Figure 4.3. Schematic simulation results of single ended 90° coupler (a) S-parameters (b) Phase difference. 4.3 180° Hybrid coupler Figure 4.4 and 4.5 shows the schematic and layout of 180° hybrid coupler designed in the frequency range of 0.7-3 GHz. The advantage of the 180° hybrid coupler is that it has even and odd modes and can be accessed separately through it sum and delta ports. The line calculator in ADS it was found that if the substrate parameter is defined and operating frequency is kept 1.85 GHz the length and width to transmission line are obtained. The length and width of the transmission lines are obtained as 0.44 mm and 0.22 mm. 0. Figure 4.4. Schematic of 180 hybrid coupler.

(42) 31. Figure 4.5. Layout of 180° hybrid coupler. Figure 4.6 shows simulation results of the single ended 180° coupler design schematic level simulation. It is noticed that there exists equal power split between output ports S 21, S31 at centre frequency. In Figure (a) the values of isolation are determined with S 41 value of -41.8 dB and at o center frequency. In figure (c) the phase of 90 phase S12 is noticed and in figure (d) it is noticed that o there exist a phase difference of 0 between S21 and S31.. (c). (d). Figure 4.6. Simulation results of 180° hybrid coupler (a) S-parameters S11, S14 (b) S-parameters S12, S13 (c) phase difference S12 (d) Phase difference between S12 - S13.

(43) 32. 4.4 180° differential coupler The 180° differential coupler is designed in the frequency range of 0.7-3 GHz frequency band. The 180° differential coupler is designed as decoupling network in between two antennas which reduces the coupling between the two MIMO antennas therefore increases the isolation between the antennas. Figure 4.7 and 4.8 shows the schematic and layout for 180° differential coupler.. Figure 4.7. Schematic of 180° differential coupler. Figure 4.8. Layout of 180o differential coupler.

(44) 33 o. o. Figure 4.9 shows the flipping layers to achieve 270 phase shift. In order to achieve 270 phase shift crossing of conductor layers from one conductor to layer to other conductor i.e. flipping from cond to cond2 layer and also simultaneously flipping other side from cond to cond2 gives 180o and remaining o o for both is 90 therefore 270 phase shift is obtained.. o. Figure 4.9. Flipping layers to achieve 270 degree phase shift. Figure 4.10 shows simulation results of 180° differential coupler layout when S11 is input then S12 and S14 acts as outputs and S13 is isolated port. It is noticed that there exists equal power split -3 dB between outputs ports S12 and S14 at the centre frequency. The isolation S13 of -39.469 dB at centre o frequency is noted in figure (a). In figure (c) it is noticed that there exist a phase difference of 0 between S21 and S41 at centre frequency.. (c). (d). Figure 4.10. Simulation results of 180° differential coupler layout (a) S-parameters S11 , S13 (b) Sparameters S12 ,S14 (c) phase difference between S12 - S41(d) Phase difference S34.

(45) 34. Figure 4.11 shows measured results of 180° differential coupler layout. The measured results were approximately same for all the S-parameters because of few losses in the prototype design process and also because of one conductor layer placed over the other there exists some variation when simulated results and measured results are compared.. Figure 4.11. Measured results of 180° differential coupler layout (a) S- parameters S21, S41 (b) Phase difference S34. 4.5 Wilkinson power divider design Power divider are passive components used for power dividing and combining used in RF, wireless and microwave communication systems. In the rapid development of communication systems power dividers of compact size, low-cost, low isolation bandwidth, insertion loss and high performance band width are required. The power divider is generally a three port device which divides equally in two output ports of 3 dB. Two wilkinson power divider are designed which are on one low frequency band with centre frequency at 1.6 GHz and other on high frequency band with centre frequency at 2.6 GHz designed, the power dividers were simulated and compared with measured results.. 4.5.1 Wilkinson low band power divider (1.6 GHz) Figure 4.12 and 4.13 shows the schematic and layout of the wilkinson power divider designed to operate at centre frequency of 1.6 GHz. The power divider is designed for equal power division o between the ports 2 and 3 and also 180 phase between the ports. The line calculator in ADS it was found that if the substrate parameter is defined and operating frequency is kept 1.6 GHz the length and width to transmission line are obtained. The length and width of the transmission lines TL3, TL4, TL 5 are found to be 6 mm and 0.42 mm .The width of the transmission line TL6 is 0.42 mm and by o varying the length up to 45.44 mm the 180 phase difference between the port 2 and port 3 is achieved..

(46) 35. Figure: 4.12. Schematic of wilkinson low band power divider at centre frequency of 1.6 GHz. Figure 4.13. Layout of wilkinson low band power divider at centre frequency of 1.6 GHz. Figure 4.14 shows the layout simulation results of the low band power divider at a centre frequency of 1.6 GHz. In figure 4.14 (a) is seen that there is an equal power division i.e., -3dB on each branch ports 2 and 3 when port 1 is input. In figure 4.14 (b) the phase difference between the ports found to be 186° which is close to the desired 180° by varying the length of one of transmission line on port the phase is varied as when both transmission lines are equal there exist 0 o phase between the output ports..

(47) 36. Figure 4.14. Layout simulation results of low band power divider at centre frequency of 1.6 GHz (a) equal power division in ports 2 and 3, (b) 180o phase difference between the ports 2 and 3. Figure 4.15 shows the measured results of the low band power divider at a centre frequency of 1.6 GHz. In figure 4.15 (a) it can be seen that there is an equal power division of approximately -3 dB between the ports 2 and 3 when port 1 is input. In figure 4.15 (b) the phase shift between the ports found to be 180°.. Figure 4.15. Measured results of low band power divider at centre frequency of 1.6 GHz (a) Equal power division in ports 2 and 3, (b) 180° phase difference between the ports 2 and 3. 4.5.2 Wilkinson high band power divider (2.6 GHz) Figure 4.16 and 4.17 shows the schematic and layout of the wilkinson power divider designed to operate at centre frequency of 2.6 GHz. The power divider is designed for equal power division between the ports 2 and 3 and also 180o phase between the ports. The length and width of the transmission lines for high band power divider considered being same as wilkinson low band power divider but length of the transmission line TL5 is varied in order to obtain 180o phase between the port 2 and 3 and at length of 28.0 mm the 180o phase is obtained..

(48) 37. Figure 4.16. Schematic diagram of wilkinson high band power divider at centre frequency of 2.6 GHz. Figure 4.17 . Layout diagram of wilkinson high band power divider at centre frequency of 2.6 GHz. Figure 4.18 shows the layout simulation results of the high band power divider at a centre frequency of 2.6 GHz. In figure 4.18 (a) is seen that there is an equal power division i.e. -3 dB on each branch ports 2 and 3 when port 1 is input. In figure 4.18 (b) the phase difference between the ports found to be 180° by varying the length of one of transmission line on one output port the phase is varied as when both transmission lines are equal there exist 0o phase difference between the output ports..

(49) 38. Figure 4.18. Layout simulation results of high band power divider at centre frequency of 2.6 GHz (a) equal power division in ports 2 and 3, (b) 180° phase difference between the ports 2 and 3. Figure 4.19 shows measured results of the high band power divider at centre frequency of 2.6 GHz. In figure 4.19 (a) it can be seen that there is an equal power division of approximately -3 dB as there was few losses in fabrication design there exist some variation between the ports 2 and 3 when port 1 is used as input. In figure 4.19 (b) the phase shift between the output ports found to be 180°.. Figure 4.19. Measured results of high band power divider at centre frequency of 2.6 GHz (a) equal power division in ports 2 and 3, (b) 180° phase difference between the ports 2 and 3.

(50) 39. ____________________________________ CHAPTER 5 5. Integrated design 5.1 Introduction Due to possibility of MIMO operation good isolation between antennas is required, so the two antennas were integrated with a fully differential coupler to enhance isolation. Figure 5.1 shows the two dipole antennas were integrated with coupler.. Figure 5.1. Antennas with coupler. Before integrating the antennas with the coupler there is some theory that needs to be considered about polarization and isolation. By conclusions from this theory it will be easy to determine how the antennas are to be placed to lessen the correlation and also to improve the isolation.. 5.2 Polarization Antenna is a transducer which converts incoming radio frequency signals into electromagnetic waves and these waves are radiated into free space. Coming to polarization, the polarization of an antenna is determined by the electric field planes. Generally an antenna radiates either linear or circular polarization [5], [32]. A linear polarized antenna is an antenna in which a plane electromagnetic wave propagates and is polarized in one single direction of propagation i.e., only one plane containing the direction of propagation [35]. Whereas in the circular polarization the plane of polarization rotated in a circle making one complete revolution during one period of the wave. If this rotation is clockwise to the direction of propagation then it is called Right Hand Circular Polarization (RHCP) and if the rotation is counter clockwise to the direction of propagation it is called as Left Hand Circular Polarization (LHCP) [34]..

(51) 40. Figure 5.2 shows polarization of antenna i.e. the linear polarized antennas (horizontal and vertical polarized antennas). A linear polarization antenna is classified into vertical and horizontal polarized antennas. A horizontal polarized antenna is an antenna where electric field vector is parallel to the earth surface and in vertical polarized antenna electric filed is perpendicular to earth surface [35], [36].. Figure 5.2. Polarization of antennas. 5.3 Isolation versus polarization There exists a simple concept for antenna to antenna communication. The horizontal polarized antennas do not communicate with the vertical polarized antennas. As per the concept of reciprocity theorem the antenna transmits and receives in exactly the same manner which means for antenna to antenna communication either both antennas should be horizontal polarized or vertical polarized. Hence a vertical polarized antenna transmits and receives with vertical polarized waves. If in a case vertical polarized antennas trying to communicate with the horizontal polarized waves there will ideally be no reception. The reception follows in a relationship cosα, where α is the angle of mismatch i.e., the difference in electric field vector orientation. If the orientation of transmitting and receiving antenna is 90o then there will be minimum power transmission since cos90 o = 0. Maximum power transfer occurs when the both are same polarized which means rotation angle is 0 o (cos0o = 1). This total concept is described by Polarization Loss Factor (PLF) [35]. PLF = cos2α. Eq. (5.1). Generally antenna isolation depends on following values Antenna type and design VSWR Distance between antennas Radiating direction Polarization.

(52) 41. Interaction between antenna systems commonly described with isolation between the antennas [36]. Isolation means low interaction between the antennas and its surroundings and is expressed in dB using S-parameters. Typically antenna isolation below -20 dB is required. Figure 5.3 shows the measurement of isolation under two possible cases i.e., both horizontal polarized and 90o phase between antennas [36]. The conclusion is that from theory it can be argued that the two antennas should have different polarization (one horizontal and one vertical) to improve the isolation between them.. Figure 5.3. Measurement of isolation under two possible cases. Figure 5.4 shows the simulated results when both are of same polarization and other of different polarization. It is noticed for a given distance say ‘d’ between the antennas, the isolation between antennas is less when both the antennas have different polarized and is more for the case 1 when both have same polarization. From the graph it is noticed that for case 1 the isolation value at 1.5 GHz frequency is -17 dB whereas in case 2 the isolation values is less than -20 dB. So finally it is observed from the simulation results the isolation between the antennas will be improved by using antennas with different polarization [37].. Figure 5.4. Simulated results when both are of same polarization and other of different polarization.

(53) 42. 5.4 Isolation versus distance The value of isolation is measured by varying distance between antennas. Figure 5.5 shows the simulation results for isolation versus distance.. Figure 5.5. Simulated results for isolation versus distance. Distance is varied and corresponding isolation between antennas is observed. From the figure 5.5. it can be observed that for d = 32 mm the isolation was much higher it reached to -27.0 dB whereas in the other two cases peak value of isolation is -24.0 dB for d = 22.0 mm, and -20.1 dB for d = 18.5 mm. By increasing the distance between antennas isolation value increases. The theory behind this is as kept on increasing the distance between two antennas it reduces negative interaction which is called coupling which occurs between two antennas; moreover power transmission between the antennas then also becomes less. This coupling reduction between antennas enables each antenna to perform at its maximum capability. The effect of change in isolation up to certain distance and after that we cannot find much further improvement in isolation [38], [39].. Table 5.1 : Isolation better than -20 dB: Distance (mm). Frequency (GHz). Bandwidth (GHz). 18.5. 1.50 – 1.60. 0.10. 22.0. 1.65 – 2.05. 0.40. 32.0. 1.85 – 2.50. 0.65.

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