Linköpingsuniversitet58183LinköpingSverige Date:2021-02-22 Examiner: LarsEriksson Supervisor: TomasUnoJonsson BillySecondNIYINZI ConvertermoduleforhighcurrentLithiumbattery LinköpingsuniversitetInstitutionenförsystemteknik(ISY)Master’sProgrammeinElectroni

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Master’s Programme in Electronics Engineering Master’s thesis 30 hp

2020

LiTH-ISY-EX–20/5355–SE

Converter module for high current Lithium battery

Billy Second NIYINZI Supervisor: Tomas Uno Jonsson

Examiner: Lars Eriksson Date: 2021-02-22

Linköpings universitet 581 83 Linköping

Sverige

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CONVERTER MODULE FOR HIGH CURRENT LITHIUM BATTERY

Billy Second Niyinzi

2021-02-22

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Abstract

Nowadays there are various types of power electronics converters available, most of them are used to convert battery with 12 V or higher. This thesis discusses and anal- yses the simulation and implementation of converter module for lower voltage/ high current (3.7V/50A) Lithium battery. The converter module consist of three major subsystems: full bridge H layout circuit that comprise of four switches to convert the unidirectional flow of current to the periodically reverse one, gate driver circuit for controlling the switching mode for the MOSFETs and the Arduino due module that is used as pulse width modulation (PWM) to control output and the switching frequencies. Different semiconductors and passive devices such as MOSFETs, diodes, capacitors and resistors were chosen and used in construction of converter and discrete gate driver. Converter model was created and examined using Simulink and the devices’ spice model and parameters provided from electronics devices manufactures. The results focused on many aspects such as circuit efficiency, power MOSFETs losses and its critical performance metrics, such as dI/dt or peak drain voltage overshoot/undershoot. The circuits implementation was done on a Printed Circuit Boards.

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Acknowledgements

I am very grateful to have performed my master’s study at Linköping University, where I met erudite professors and collaborative colleagues who helped me a lot during my masters journey. I would like to take a moment to express my gratitude to all people who supported me during this project. Let me begin by expressing my gratitude to my supervisor Tomas Uno Jonsson for his valuable support, guidance, constructive ideas and useful advice throughout this thesis. I would like to thank my examiner Professor Lars Eriksson for auditing my thesis and providing me insightful feedback. I would also like to express my thanks to Mr. Arvind Balachandran for his willingness to help and wonderful advices. Finally, I must reserve the most special appreciation to my family for always believing in me, especial my lovely, patient wife for her support and encouragement during my master’s at Linköping University.

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List of Figures

2.1 Classification of power converter . . . 3

2.2 H-Bridge converter . . . 4

2.3 Output wave signals comparison for H-bridge DC/AC converter . . . . 5

2.4 DC motor direction control . . . 6

2.5 Circuit structure of a modular multilevel converter with half-bridge SMs. 7 2.6 Half bridge submodule circuit structure. . . 8

2.7 (a) Perspective [4] and (b) Cross-sectional view of an N-channel power MOSFET, (c) MOSFET symbol . . . 10

2.8 Operation principle of N-channel MOSFET . . . 10

2.9 N-Channel E-MOSFET [5](a) Transfer characteristics (b) Output characteristics . . . 11

2.10 MOSFET in OFF mode and equivalent switch circuit . . . 12

2.11 MOSFET in ON mode and equivalent switch circuit . . . 13

2.12 (a) MOSFET equivalent circuit with parasitic components that have the largest impact on switching [9], (b) MOSFET equivalent gate circuit 13 2.13 Switch on transients of the MOSFET [10] . . . 14

2.14 Switch off transients of the MOSFET [10] . . . 16

2.15 VGS and VDS waveforms [10] . . . 17

2.16 Path of current during the switching of MOSFET. . . 17

2.17 Blanking time and output voltage losses . . . 18

2.18 Typical linear hard-switching waveforms representation for power MOS- FET switching transition and power losses . . . 19

2.19 (a) MOSFET Steady-State Thermal Resistance Model, (b)Electrical model [4] . . . 20

2.20 (a) Constant duty cycle and (b) varying duty cycle . . . 21

2.21 Voltage waveform of bipolar modulation . . . 22

2.22 Unipolar PWM modulation Voltage waveform . . . 23

2.23 Half bridge gate driver block diagram . . . 24

2.24 Lower side MOSFET gate driver [13] . . . 24

2.25 Lower side MOSFET gate driver with Resistance for limiting shoot through . . . 25

2.26 High side MOSFET gate driver [13] . . . 25

2.27 Simple structure of high side gate driver circuit with bootstrap circuit. (a) Bootstrap capacitor charging, (b) Bootstrap capacitor discharging. . 26

3.1 Design topology with gate drivers . . . 28 vii

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4.1 Simulink model of Half-bridge converter . . . 32

4.2 Half bridge converter with gate driver . . . 32

4.3 Simulink model for MOSFET gate control driver block . . . 33

4.4 Simulink model PWM block . . . 34

4.5 Input vs output current and voltage waveform from simulation . . . 36

4.6 Gate-source voltage charging and discharging input capacitance . . . . 36

4.7 Waveform representation for pulse digital 1 and output current rising of the power MOSFET . . . 37

4.8 Waveform representation for 0 digital pulse and output current falling of the power MOSFET . . . 37

4.9 Waveform representation for drain-source voltage rising and falling . . 38

4.10 Gate driver without clamping MOSFET . . . 39

4.11 Gate input capacitance charging voltage graph . . . 40

4.12 Rising current graph with gate drive without clamping MOSFET . . . 40

4.13 Current falling graph with gate drive without clamping MOSFET . . . 41

4.14 Drain source voltage VDS falling graph. . . 41

4.15 Drain source voltage VDS rising graph . . . 41

5.1 PCB design schematic diagram . . . 44

5.2 Q1 Gate driver schematic with components label for SC1 . . . 44

5.3 PCB top layer layout . . . 44

5.4 PCB component labeling and layout . . . 45

5.5 PCB Final full bridge convertor with components assembled . . . 46

5.6 Testbench block diagram . . . 46

5.7 Test bench setup with oscilloscope and measurements tools . . . 47

5.8 VGS Switching off transients for power MOSFET . . . 47

5.9 VGS charging input capacitance of power MOSFET . . . 49

5.10 Drain current ID rising . . . 49

5.11 Drain Source voltage VDS rising. . . 49

5.12 Output voltage measured on terminals of resistance load and output current. . . 50

viii LIST OF FIGURES

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List of Tables

2.1 Allowed switching states . . . 4

2.2 DC motor control switching states . . . 7

2.3 MOSFET Junction capacitance . . . 14

2.4 MOSFET Junction capacitance . . . 14

3.1 Design Specification . . . 27

3.2 Power transistors comparison [14] . . . 29

3.3 Components number and rating of used devices . . . 30

4.1 Gate driver power supply characteristics. . . 33

5.1 Gate driver power supply characteristics. . . 45

5.2 Measured and simulated parameters comparison. . . 48

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x LIST OF TABLES

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List of Acronyms and Abbreviations

AC : Alternating Current

DC : Direct Current

P V : Photo-voltaic

U P S : Uninterrupted Power Supply P W M : Pulse Width Modulation

SP W M : Sinusoidal Pulse Width Modulation T HD : Total Harmonic distorision

P CB : printed circuit board

M OSF ET : Metal Oxide Semiconductor Field Effect Transistor

EM OSF ET : Enhanced Metal Oxide Semiconductor Field Effect Transistor N − M OSF ET : NChannel Metal Oxide Semiconductor Field Effect Transistor P − M OSF ET : PChannel Metal Oxide Semiconductor Field Effect Transistor IGBT : Insulated Gate Bipolar Transistor

BJ T : Bipolar Junction Transistor GU I : Graphic User Interface

SOA : Safe Operating Area

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xii LIST OF TABLES

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Chapter. 1

Introduction

Today the world is (worried) concerned about global warming caused mainly due to the use of fossil fuel in energy generation. To reduce and solve atmospheric pollution and meet the increasing energy demand, renewable energy resources and use batteries for energy storage is one of the solutions. The electrical power stored in batteries requires control and conversion to meet modern technology power supply requirement.

Then the converters are needed. Converters take DC power from battery as input and convert it into required output power supply. The design of converter module for high current lithium battery 3.7V / 50 A, involve the design of high efficiency switching power stage and gate control stage in order to achieve a suitable conversion. The power stage circuits are the one in charge of conversion and is based on H-bridge switching converter topology. The lower voltage (3.7V) specification for our converter challenged us to design also its appropriate discrete gate driver, as the available standard single chip gate driver circuits was only found for gate voltages above 8-10V. Then discrete two stage gate drive have been designed to drive power MOSFETs. The first stage is working as level shifter and the second as buffer to produce enough current driver input for the gate of power MOSFETs. Arduino due microcontroller have been used to generating pulse width modulation.

1.1 Motivation

Nowadays the demand of clean and renewable energy consumer devices (electrical appliance) are increasing. For example, EVs, portable charges. . . . These devices require a certain amplitude and frequency of voltage to run properly or to be controlled (speed or direction). As many sources of electrical power produce a certain constant and fixed amount of voltage like a battery, then in order to be used for different devices the converters are required for adjusting the source voltage to the devices’ need.

1.2 Aims (Purpose)

The aims of this project are to design, simulate and implement a converter capable to convert a lower voltage high current DC to AC, output voltage of 3 t0 4V/50A AC from a high current lithium battery of 3.7V. This converter can be used for different

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applications such as automotive drive systems, photovoltaics (PV) panel systems or energy convert from energy storage system like Uninterrupted power supply (UPS).

1.3 Methodology (Scope)

To achieve the aim and objectives of this works, the following steps are involved:

the first step is to analyze the converters theoretically into their parts and working principles by considering topologies, components and modulation.

After, the second step will deal with the performance parameters specification in order to select the appropriate components. Therefore, several calculations and simulations will be necessary.

Once theoretical simulation and concrete components have been chosen, the converter module will be implemented on PCB in order to obtain the real circuit module.

Also, the final real converter module will be simulated by using a computer software and Arduino due as microcontroller to generate PWM, to be able to have a better idea on expected results.

Finally, the comparison between calculation, theoretical simulation and implemented converter will be done and resume of the objectives achieved, as well as con- clusion and recommendations.

2 CHAPTER 1. INTRODUCTION

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Chapter. 2

Theoretical background

2.1 Converter

Power converter is an electronic circuit for energy conversion, which converts electrical energy form of the supply into the desired energy form optimized for the specific load.

Converters can be classified into four major types based on the input and output voltages namely the AC to AC cyclo-converter (frequency changer) or transformer, the AC to DC converter known as the rectifier, the DC to DC voltage or current converter known as chopper, and the DC to AC inverter [1]. Converter uses nonlinear components as switches such as transistors and diodes, and linear reactive components such as the capacitors, inductors and, transformers for intermediate energy storage as well as current and voltage filtering

Figure 2.1: Classification of power converter

There are many different converter topologies: Based on input and output peak voltage amplitude we have step up (boost) Converter, step down (buck) converter and buck-boost converter. Based on their input energy storage method (or input feeding) we found current source converter and voltage source converter. Based on the switching devices and legs number, converter can be classified as half bridge converter and full bridge converter (called also H-Bridge converter) which will be used in this project due to its high efficiency.

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2.1.1 Full bridge converter

A full H-bridge converter topology based on power electronic switches (power MOS- FET in this project) is shown in Figure 2.2. The freewheeling diodes used to permit flow of current even if all MOSFET switches are not conducting, these diodes also permit lagging current to flow in inductive loads.

Figure 2.2: H-Bridge converter

In order to avoid the short circuit of the input supply voltage Vin, the switching mode of the four MOSFET’s switches (S1, S2, S3, S4) should be controlled so that either S1 or S2 is on(closed), but never at the same time and either S3 or S4 is on but never at the same time. As most of the inverter use pulse width modulation (PMW) technique to control the switching period. There are two type of switching mode known as bipolar switching and unipolar switching or three levels switching.

In bipolar scheme the output voltage Vout swing between +Vin and −Vin, whereas the output voltage Voutof three levels system (unipolar) swing between +Vin, 0, −Vin. The following combination for the MOSFET’s switches should be applied and dead-time or blanking time should be implemented in order to avoid short circuit.

Table 2.1: Allowed switching states

4 CHAPTER 2. THEORETICAL BACKGROUND

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Design and implementation of converter module for high current Lithium battery

2.1.2 H-Bridge converter application

H-bridge converter is often used in two major applications:

I. DC/AC converter

H-bridge converter designed to converter DC input power supply to AC output power supply. DC /AC converters are general classified into three main type based on the shape of their output wave form voltage:

i. Square wave DC/AC converter ii. Modified sine-wave DC/AC converter iii. Pure sine-wave DC/AC converter

Figure 2.3: Output wave signals comparison for H-bridge DC/AC converter

i. Square wave DC/AC converter

Square wave DC-AC Converter is considered to be the simplest class in design and implementation of DC-AC Converter which converts DC signal to an AC. However, the drawback of this simplest is that the power dissipation is considerably high due to its lot of amount of total harmonic distortion (THD) of its output waveform. Other disadvantage of square wave DC/AC is that it cannot regulate its output AC voltage amplitude when the input DC change significantly. This can cause some load to suddenly fail.

ii. Modified sine-wave DC/AC converter

Modified sine wave DC/AC converter wave form is between sine wave and square wave form. Its output wave form is similar to the square wave except that it has three steps known as positive, zero and negative (it has a step or dead space between positive and negative square waves), while square wave has only positive and negative. Before switching from positive to negative or vis versa it goes to zero for some time delay.

THD is reduced due to this three steps and it is possible to regulate average output voltage level by controlling the pulse shape of the square wave which can be made

CHAPTER 2. THEORETICAL BACKGROUND 5

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tall and narrow or short as the battery input voltage change from high to lower. This result on improving better load compatibility and performance.

iii. Pure sine-wave DC/AC converter

Pure or true sine wave DC/AC converter is the most complex in design and implementation. It produces the output waveform which is closely matches to the from of the grid supply system. Pure sine wave converters have considerable low THD, that is why it can be used run more sensitive AC loads. In pure sine wave DC/AC converter, a high frequencies PWM is used to produce a hundreds of positive and negative pulses during each AC cycle, then this pulses are filtered to give smooth output sine wave shape. Output voltage and frequency of pure sine wave converter can be also controlled to give better load compatibility and performance.

II. DC Motor load controller

A full bridge converter is used for ac motor control as whereas dc motor control. In AC motor control full bridge converter used to control its speed, while in dc motor H-bridge can be used to control speed, position and direction. The DC motor direction control is illustrated in the Figure 2.4 where the switching MOSFETs are usually used to reverse the motor’s polarity / direction, but can also be used to ’ brake ’ the motor as it can be suddenly stopped when the motor’s terminals are shortened, or motor’s terminals can be disconnected to the circuits and lets the motor free run to a stop.

Table 2.2 shows the possible combination of DC motor direction and stopping control.

Figure 2.4: DC motor direction control

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Table 2.2: DC motor control switching states

2.1.3 Modular Multilevel Converter MMC

The Modular Multilevel Converter (MMC) is an advanced converter topology with scalable technology that makes high voltage and power capability possible. The MMC is built up by identical Cascaded connection of submodules (SMs) or cells (made up of half-bridges or full bridges), but individually controllable. Therefore, the converter can act as a controllable voltage source. Normally MMC topology uses distributed dc- capacitor in converter module. An example of MMC 3-phase topology implemented with half bridge is shown on the following Figure 2.5.

Figure 2.5: Circuit structure of a modular multilevel converter with half-bridge SMs.

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In a 3-phase MMC topology, each of the phase unit (leg) consists of two arms (upper and lower), and each arm consist of N half bridge submodules connected in series.

Half-bridge submodule is composed of two semiconductors switches (MOSFETs in our example with antiparallel diodes) and a capacitor. Normally only one MOSFET is switched on at a given instant in time, which allows submodule either to insert capacitor in the arm or to bypass it [2]. As capacitor may be considered as a DC voltage source, insertion or bypass it will result in either a finite or zero DC voltage contribution in the arm. Three possible switching states ca be defined for half bridge module [3].

• Insertion state: Q1 is on, and Q2 is off. Therefore, the submodule output voltage VSM, equal to the capacitor voltage VC.

• Bypass state: Q1 is off, and Q2 is on. The submodule output voltage VSM is zero, and the capacitor voltage remain constant.

• Blocking state: Both MOSFETs (Q1 and Q2) are off, current can only flow through the freewheeling diodes.

Figure 2.6: Half bridge submodule circuit structure.

For a constant module capacitor voltage, converter should have balanced input and output energy i.e., DC power should be equal to AC power. When power balance does not apply the module capacitor will be charged or discharged depending of direction arm current.

Main application of MMC are:

• High voltage power, high power transmission system (HVDC).

• Variable speed motor driver (VSD).

• MMC can be also used as either source or sink of reactive AC power to an electricity network or as static synchronous compensator (STATCOM).

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2.2 Power MOSFET

MOSFET stand for Metal Oxide Semiconductor Field Effect Transistor. The MOS- FET transistor semiconductor device commonly used in digital design (switching) and amplification of power signals (current and voltage) in electronic devices. MOSFET is a voltage control device as output current is not dependent of input current, rather it is the gate-source voltage that controls the drain current. The MOSFET is four terminals device called gate (G), drain (D), source (S) and body (B) which serves to modulate the MOSFET characteristics and parameters. The Body electrode is connected directly to the substrate and is frequently connected to the source terminal so making MOSFET a three terminals device transistor. Region between drain and source layers is called channel and has two important geometries characteristic know as channel length (L) and Channel width (W) which used in calculation of MOSFET’s parameters.

2.2.1 Basic structure of power MOSFET

A Power MOSFET physical structure basically has the vertically oriented four-layers of alternating p-type and n-type doping as shown on Figure 2.7, the channel length is still horizontal like in lower power MOSFET, but it is much shorter to minimize the gate capacitance. Furthermore, the current flow between channel and drain is vertical while in standard lower power MOSFET, the current flow horizontal.

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Figure 2.7: (a) Perspective [4] and (b) Cross-sectional view of an N-channel power MOSFET, (c) MOSFET symbol

The additional layer in Power MOSFET between n-plus drain layer and p-substrate is called drift region and determines the voltage-blocking capability of the devices.

The vertical nature of power MOSFET has also an advantage for facilitate parallel connection of many devices to increase current handling capacity without suffering from current hogging and thermal runway.

2.2.2 Working principle

The MOSFET works by electrically inducing a channel along which charge carriers flow (electrons or holes), so that current flow between drain and source can be controlled.

For example, when we apply a positive voltage on the gate of N-MOS, an electric field will be created within the P-type substrate, which will push the positively charged carriers in it away from the gate, while negative charged carriers within the p-substrate are attracted to the positive gate and accumulate close to the surface of the substrate below the gate because these electrons cannot pass through the silicon dioxide isolated layer to the gate. The electrons reach channel is formed between source and drain.

Now, if a voltage is applied between the drain and source, the current flows freely between them through the channel and the flow of current is controlled with gate voltage by controlling the conductivity of channel (electrons in the channel).

Figure 2.8: Operation principle of N-channel MOSFET

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Figure 2.8 show the operation of N-channel E-MOSFET, where gate is connected to positive voltage, if gate-source voltage (VGS) is high than threshold voltage (VT H) there will be induced channel between drain and source, and when the drain-source voltage (VDS) is positive the current ID will flow from drain to source through induced channel. If we increase VGS, more amount of negative carriers is increased in induced channel result in increasing of current ID.

2.2.3 Characteristics of MOSFET

N-type E-MOSFET’s transfer characteristic or transconductance curve is shown on the Figure 2.9 (a) Transfer characteristic is the graphic representation of drain current ID in function of VGS when VDS is constant. The characteristic curve also show that, ideal there is no drain current until VGSreaches a certain non-zero MOSFET parameter called threshold voltage (VGS(th)). On Figure 2.9 (b) represent output characteristic of N-type E-MOSFET where the drain current ID versus drain source voltage VDS curves are represented graphically at different gate source voltage VGS.

Figure 2.9: N-Channel E-MOSFET [5](a) Transfer characteristics (b) Output characteristics

MOSFET has three operation regions as shown in Figure 2.9 depending upon the relative voltages of its terminals, as shown on output characteristics:

1. Cut-off region: MOSFET is said to operate in cut-off region when the absolute gate-source voltage |VGS| is less than absolute threshold voltage |Vth|. In this mode there is no drain current flow through the MOSFET, and it behaves like open-circuit or Off.

2. Linear or resistive region: MOSFET is said to operate in linear mode when

|V_GS| > |V_{T H}| and VDS < V_DS,sat. The drain current ID increases linearly with increasing drain-source voltage VDS. In this region, the device acts as a resistor with almost a constant on-resistance, (RDS(on)) defined by VDS/I_D.This region is mostly used when MOSFETs are working as amplifiers. It can also be used as switching MOSFET to get lowest loss.

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3. Saturation region: If |VGS| > |V_{T H}|, there is a particular level of VDS voltage called pinch-off voltage or VDS,sat, beyond which, increasing drain-source voltage VDS appears to have no effect on drain current. Under this condition, ID is still dependent on VGS, since increased VGS increases the inversion layer density.

When MOSFET is used as switch, it work only on two regions: cut-off region for Off position and linear region for On position. Current ID trough the MOSFET is a function of the gate-source voltage VGS and determined by equation 2.1:

I_D = K.(V_GS− V_{T H})² = g_{f s}.(V_GS− V_{T H}) (2.1) Where:

• K is a constant and depends on MOSFET’s parameters. K can be determined from the datasheet by taking the specified value of ID, called ID(on), at the given value of VGS and substituting the values into equation.

• and gf s is the current gain or transconductance of the device and is defined as the change in drain current divided by the change in gate voltage for a constant drain voltage [6], [7].:

g_{f s} = dI_D

dV_GS|VDS=constant (2.2)

2.2.4 MOSFET as electronics switch

In power converter we use MOSFET as electronic switches where they operate in cut- off or linear regions. In this project we chose n-type E-MOSFET because it requires positive voltage to switch “ON” (closed) and zero voltage to switch “OFF” (open) making it easily to be used in converter. The basic function of MOSFET working as switch is to control the flow of drain current ID by gate source voltage VGS. When the gate-source voltage is less than threshold voltage VGS < V_th, there is no drain current I_D = 0, because MOSFET is not conducting or is in OFF mode and act as an open switch. Equivalent circuit is shown on the Figure 2.10.

Figure 2.10: MOSFET in OFF mode and equivalent switch circuit

Then the MOSFET will conduct (ON mode) when it is working in linear region. In this mode the gate source voltage VGS is much greater than threshold voltage Vth, so

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that the drain source resistance is decreased from very high OFF resistance to the ON resistance RDS(ON ) very lower so that the MOSFET effectively acting as short circuit, and the drain current is reaching its maximum. [8].

Figure 2.11: MOSFET in ON mode and equivalent switch circuit

2.2.5 Switching properties of the MOSFET

In this project we will use MOSFET as switch, that why we need to analyze the behaviors of MOSFET in switching mode. The switching process and time of the MOSFET are influenced and limited by its internal capacitance and the gate drive circuit’s internal impedance. The following Figure 2.12 (a) shows the power MOSFET equivalent circuit with parasitic components that have the largest impact on switching.

In addition, it includes a body diode, which is due to the physical structure of the device. In other side, the equivalent gate circuit with gate resistance and parasitic gate capacitances on Figure 2.12 (b) which demonstrate gate current distribution for charging input capacitances.

Figure 2.12: (a) MOSFET equivalent circuit with parasitic components that have the largest impact on switching [9], (b) MOSFET equivalent gate circuit

• Turn on behavior of MOSFET:

If a gate terminal is connected to a positive supply voltage, the gate source capacitance CGS starts being charged. At the same time gate drain capacitance Cgd is

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discharged by the same amount of voltage change as the gate voltage increase. When gate-source voltage VGS reaches threshold voltage Vth, drain current ID start rising.

As CGS continue to charge, gate source voltage VGS and drain current ID continue to rise proportionally until CGS is full charged and the current reaches its maximum. At that moment gate-source voltage VGS stays constant while the drain-source voltage starts to decrease and gate-drain capacitor CGD start to be charged.

Figure 2.13: Switch on transients of the MOSFET [10]

In order to simplify the equations, from manufactures data-sheets, parameters are defined as input capacitance (Ciss), reverse transfer capacitance (Crss), output capacitance (Coss) and gate resistance (RG). These parameters are detailed in the following table.

Table 2.3: MOSFET Junction capacitance

Parameters Explanation Interpretation

R_G R_Gint+ R_Gext Gate resistance

C_iss C_GS+ C_GD Input capacitance

C_rss C_GD Feedback capacitance

C_oss C_DS+ C_GD Output capacitance

Table 2.4: MOSFET Junction capacitance

Then we define time t1 as the time as the delay it takes to charge the input capac-

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itance from zero voltage to threshold voltage.

t1 = −RG× Ciss× ln(1 −V_{T H}

V_in ) (2.3)

Time, t2 is the time required to charge input capacitance Ciss to miller plateau voltage (VGP).

t₂ = −R_G× C_iss× ln(1 −V_GP Vin

) (2.4)

Note that at given ID current, Miller Plateau voltage VGP can be calculated using equation 2.1 as follow

V_GP = V_GS = I_D

g_{f s} + V_{T H} (2.5)

Then the calculation of time t3 is more tricky as CGD changes with VDS and gate- source voltage VGSstays constant at miller Plateau voltage VGP, and all of gate current goes to discharge this feedback capacitance CGD from VDS to almost zero, when MOS- FET is full on. So that t3 can be determined using dV/dt capability, which is defined as rate of rise of drain-source voltage allowable. If this rate is exceeded then the voltage across the gate source terminals may become higher than the threshold voltage, forcing the MOSFET into current conduction mode. And we can calculate it by consider the feedback current (IG) flowing from drain to gate though CGD and RG.

I_G = V_in− V_GS

R_G = C_rss× dV

dt (2.6)

Then t3 is calculated as

t₃ = C_rss× V_DD − V_DS(on)

I_G (2.7)

If we substitute equation 2.6 into equation 2.7 we gate

t3 = Crss× RG× V_DD − V_DS(on)

V_in− V_GS (2.8)

• Turn-off behavior of MOSFET:

When the input voltage (gate-source voltage) is reset to zero or to negative power supply, input capacitance Ciss starts to discharge through gate resistance RG until it reaches miller plateau voltage VGP where the MOSFET output voltage (drain-source voltage VDS) start to increase. At that moment gate-source voltage VGS stays at VGP as the gate current is flowing through gate-drain capacitance CGD and discharge it while charge in gate-source capacitance CGS remain constant. When drain-source voltage V_DS reaches its maximum values, capacitance CGS start discharging with decreasing gate source voltage VGS and drain current ID decrease proportionally.

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Figure 2.14: Switch off transients of the MOSFET [10]

Then the switch off transient times can be calculated using the following formulas

t₄ = R_G× C_iss× ln(V_GS

V_GP) (2.9)

t₅ = R_G× C_rss× V_DS

V_GP (2.10)

t₆ = R_G× C_iss× ln(V_GP

V_th ) (2.11)

As you can see from the Figure 2.13 and Figure 2.14, the combination of all transients delay are used to estimate rise and fall times of voltage and current for the MOSFET, we use them also to determine turn-on and turn-off delays of MOSFET.

• Current rise time tri: tri = t₂− t₁

• Voltage fall time tvf: tvf = t3

• Voltage fall time tvf: tvf = t₃

• Voltage rise time tvr: tvr = t₅

• Current fall time tif: tif = t₆

According to MOSFET’s data sheet, we define the turn-on delay, td(on), as the time interval measured between 10% of VGS rising from zero and 90% of VDS falling from rated voltage. Similarly, turn-off delay, td(of f ), is is defined as the time interval measured between 10% of VDS rising from zero and 90% of VGS falling from rated voltage [11].

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Figure 2.15: VGS and VDS waveforms [10]

2.2.6 Overshoot voltage

Voltage overshoot refers to the transitory values of voltage that exceeds its final (steady state) value during its rising state (as seen on VDS of Figure 2.14). The dominating effect on the voltage overshoot at turn-off and voltage dip at turn-off is given by the voltage drop of the parasitic inductance to the dc-side, shown in Figure 2.16. At turn- on of MOSFET Q1 the current from the dc-side flows according to the red arrow. This gives a voltage drop of equal to LS× dI/dt reducing VDS of MOSFET Q1. At turn-off of Q1 the current flows in the opposite direction of the red arrow giving a voltage rise of VDS1.

Figure 2.16: Path of current during the switching of MOSFET.

2.2.7 Blanking time

Ideally, the switching in MOSFET is instantaneous i.e. MOSFET switches ON or OFF immediately when it receives appropriate gate signal. Such behavior makes it possible to avoid overlapping for two complementary ideal MOSFETs in one leg.

However, in practical full-bridge converter, there is a possibility of short circuits across the input DC bus as it is possible for one MOSFET to turn on while the other is still in the process of turning off. Since in reality a MOSFET does not instantly go from ON to OFF, or OFF to ON, but instead has a gate charge and discharge time, a dead time (or blanking time) should be introduced between the instant at which one MOSFET

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turns OFF and the instant at which the other MOSFET in the same leg turns ON.

The dead time is the period at which neither the high nor the lower MOSFET of the same leg can be conducting.

Figure 2.17: Blanking time and output voltage losses

Blanking time has drawback of causing distortions of the output voltage of the converter, known as blanking time effects or dead time effects. The more dead time you have the less efficient your converter will be, and if the dead time is too short converter suffer the shoot through problem which can damage MOSFET due to high current.

2.2.8 MOSFET power losses

In converters where power MOSFETs operate in switch mode, power consumption is generated by three factors known as conduction, switching and leakage (leakage is normally neglected).

Typical linear hard-switching waveforms are illustrated in Figure 2.18. where waveform of the MOSFET gate-source voltage represented by VGS, VDS represents the voltage across the drain-to-source and ID represents the MOSFET current waveform.

Switching loss occurs at turn on when VGS transitions from threshold voltage VT H

to Miller Plateau voltage VP L (when VDS = 0) and at turn off when VGS transitions from VP L down VT H (when VDS begins to ramp up). Conduction losses occur when MOSFET is in steady state on condition and is due to the MOSFET drain-sourse on resistance RDS(on.)

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Figure 2.18: Typical linear hard-switching waveforms representation for power MOSFET switching transition and power losses

As power MOSFET mainly has an antiparallel diode, the total losses are composed with MOSFET losses plus losses in antiparallel diode [12].

• Conduction losses

Conduction losses in power MOSFET are generated when MOSFET is in on state and the maximum current ID is flowing between drain and source. Conduction losses are calculated with the rms value of the on-state current (the drain current rms is defined by the duty cycle) and MOSFET on-resistance RDS(ON ).

PCM = R_(DS(on)).I_Drms² = R_(DS(on)).(IOut× D)² (2.12) Conduction losses in antiparallel diode is done using diode rms forward current which defined by the blanking time. These losses can be calculated using the following formula [12]:

P_CD = U_D0.I_{F av}+ R_D.I_{F rms}² (2.13)

• Switching losses

Switching losses in power MOSFET occurs when it switches from one state to another and can be approximately calculated by using the following formula:

P_sw = 1

2.V_DC.I_o.f_sw× (t_sw(0n)+ t_{sw(of f )}) (2.14) or

P_sw = 1

2.V_DC.I_o.f_sw× (t_ri+ t_{f v}+ t_rv+ t_{f i}) (2.15)

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Where output current Io can be calculated as Io = I_Lav for DC to DC converter and I0 = ⁽²

√2)

π × I_Lrms for DC/AC converter.

Switching losses in antiparallel diode ca be approximately calculated using equation 2.16, [12]:

P_{SW D} = E_onD.f_sw = 1

4f_sw.Q_rr.U_Drr (2.16) Normally leakage losses is very small and can be neglected, then the total losses can be approximate as the sum of conduction and switching losses.

P_loss = P_CM + P_{SW M} + P_CD + P_{SW D} (2.17)

2.2.9 MOSFET thermal behavior

One of important losses on MOSFET is conduction losses which occurs when device is on conduction state (on state). This loss is proportional to the drain source on resistance (RDS(on)). Since some MOSFET’s structure components are linked to the temperature, also the total RDS(on) value is proportional with the temperature as result the power losses are due to the Joule’s effect in the structure. Then, the power dissipation in power semiconductors normally increases with the internal temperature.

In order to ensure maximum reliability, device manufactures specify on datasheet, the maximum operating temperature. For semiconductor devices the maximum allowable power dissipation that will raise the die temperature to the maximum allowable when the case is held at 25⁰C temperature is calculated by the following formula:

Pd = Tjmax− 25

R_θjc (2.18)

Where, Pd : Maximum power dissipated in the junction, Tjmax is the maximum allowable p-n junction temperature in device (normally 150⁰C or 175⁰C) and Rθjc is the junction to case thermal impedance of the device. The calculation of this power dissipation is used to ensure that the devices is working within their specification in order to achieve optimal performance and reliability. In case that the MOSFET power dissipation is generate high temperature than the allowable, the additional external cooling system must be used (Heat sink or fan). Then thermal analysis of any semiconductor device can be done by modeling the different junctions and temperatures as electrical circuits.

Figure 2.19: (a) MOSFET Steady-State Thermal Resistance Model, (b)Electrical model [4]

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When using electrical model to analyze the heat generation in MOSFET, the temperature is considered as voltage, power is considered as current and thermal resistance Rθ is considered like ohmic resistance.

Then the junction ambient can be calculated as all series thermal resistance and can calculated as follow:

Rθja= T_j− T_a

P = Rθjc+ Rθcs+ Rθsa (2.19) From this equation one can write expression of heating power P:

P = T_J − T_a

R_θjc+ R_θcs+ R_θsa (2.20)

so that in order to keep heating power dissipation in MOSFET as lower as possible, if we suppose ambient temperature is constant then Rθsa should be chosen to ensure stability.

2.3 Pulse width modulation (PWM)

In converter the switching period is controlled by pulse width modulation (PWM) where a switching frequency of a triangle wave form is compared with the frequency of control signal voltage Vcontrol. The basics function of PWM in converter module is to control the duty cycle. The duty cycle defines the amount of time the signal is in a high (on) state related to the its period.

D = T_on

T (2.21)

In converter design we can use PWM with fixed duty cycle when the control signal is constant (DC signal) or we can have a varying duty cycle when we use an AC control signal where in general we use sinusoidal signal and we call it sinusoidal PWM (SPWM).

Figure 2.20: (a) Constant duty cycle and (b) varying duty cycle

We have two important parameters to consider in PWM, one is the frequency modulation ratio (mf) which is the ratio of switching frequency with frequency of control signal

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m_f = f_sw

f_cont (2.22)

the second parameter is the amplitude modulation ratio which is often called the modulation index, and is the ratio of the peak amplitude of the control signal with the peak amplitude of the switching signal

m_a= V_cont

V_tri (2.23)

2.3.1 Bipolar PWM method

In bipolar modulation method the output signal oscillates between two value which are negative and positive hence its name. In this method, both legs of a full bridge converter are controlled simultaneously, the diagonally opposite MOSFET are switched on or off at the same time. Each half cycle of the control signal generates both positive and negative signals in the output so that only one control signal are required. Bipolar PWM is easier to implement but it has high total harmonic distortion (THD), in order to be able to filter out harmonic signals, a large filter is required, and a high switching frequency can be used with drawback of increasing switching losses. In bipolar modulation the output voltage (VA and VB) of each leg of H-bridge share the common ground, then the total output voltage (VAB) is the difference between legs voltages as shown in Figure 2.21.

Figure 2.21: Voltage waveform of bipolar modulation

This modulation has an important advantage in conversion of the constant voltage to ground (common mode voltage), which make bipolar modulation better suited for transformer-less full bridge PV-inverter.

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2.3.2 Unipolar PWM

Unlike bipolar modulation, when unipolar modulation is implemented in the circuit, there are three voltage level in the output known as positive, zero and negative. MOS- FETs for each H-bridge converter leg are controlled independently of the other. As the MOSFETs must be complementary in performance, both legs of the converter share the same switching signal. Furthermore, two sinusoidal control voltage with same frequency and magnitude, and 180^◦ degree phase shift are required. The advantages of unipolar is that harmonics that are odd multiples of switching frequency are cancelled (mf, mf+2, mf-2 etc for odd mf). Only even multiples are obtained as 2*mf-1, 2*mf+1, 2*mf-3, 2*mf+3 etc. Therefore, filter requirement is less at the output. In unipolar modulation, only one sign (positive or negative) appears at output on each half cycle of the control signals, hence its name.

Figure 2.22: Unipolar PWM modulation Voltage waveform

2.4 Gate control driver

A MOSFET gate driver is a device which generate the requirement current to charge and discharge gate input capacitance Cissfor the power MOSFET or in other words it is interface between control signals and power MOSFET. As the control signal generated from PWM is a digital or analog with lower current, a power amplifier that can accept lower current and amplify it to the appropriate high-current gate driver for power MOSFET is required, since its name. In general, power MOSFET gate drive is composed of a level shift combined with an amplifier. The advantage of the gate driver is to improve the switching time delay and providing galvanic isolation (Protection against unwanted current from flowing between power MOSFET and PWM generator) in the circuit.

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Figure 2.23: Half bridge gate driver block diagram

Gate drivers can be provided on-chip as integrated circuit, or they can be designed as a separate module. There are two type of gate driver known as lower-side and high-side gate driver.

2.4.1 Lower-side gate driver

Lower side gate drive is the driver designed to generate the requirement current for lower side power MOSFET of one half of the H-bridge converter.

Figure 2.24: Lower side MOSFET gate driver [13]

Lower side driver is composed by two stages, first one is level shifter that is used increase the digital pulse to the voltage level required for power MOSFET, the second stage is the buffer that can provide enough current to quick charge and discharge input capacitor of power MOSFET. Current buffer can be designed by the complementary MOSFET (CMOS), with low on resistance (RDS(on)).CMOS is a PMOS-NMOS combination, and both are controlled from a common signal since their gate-source thresholds are inverted with respect to each other. Because of that, during transition of gate voltage from lower to high both PMOS and NMOS transistors conduct simultaneously which create shoot through. Then a resistance is added on the output as shown on Figure 2.25 , to limit turn-on and turn-off current as well as the shoot through current when PMOS and NMOS conduct simultaneously

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Figure 2.25: Lower side MOSFET gate driver with Resistance for limiting shoot through

2.4.2 High-side gate driver

High-side gate driver is more complex than lower-side, the issue with turning on high- side N-MOSFET is referencing. Because the source terminal of high-side N-MOSFET is not at common ground, it will be floating depending on the status of the switching MOSFET. If the high side MOSFET is on, its source terminal will be at almost Vbat; if the high N-MOSFET is off, the source will be near zero through low side N-MOSFET (as the MOSFET are complementary). Thus, to turn the high-side N-MOSFET on, the gate driver needs to create an effective gate voltage that is higher than the input voltage. One approach that solves this challenge is the addition of some kind of charge- pump to generate gate voltage higher than battery voltage, mostly in bootstrapped configuration as shown on the following figure 2.26.

Figure 2.26: High side MOSFET gate driver [13]

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Bootstrap circuits

A bootstrap circuit is a circuit used to step up gate voltage of high side n-channel MOSFET. When n-channel MOSFETs are used as power switches on both sides of converter (high side and lower side), turning on high side MOSFET requires a high gate voltage because its source terminal is connected to the load while its drain is connected to the power supply. A bootstrap is a charge pump circuits compose with a diode, and a capacitor used to step up a gate voltage to the voltage high than the source voltage in order to completely turn on high side n-channel MOSFET.

Figure 2.27: Simple structure of high side gate driver circuit with bootstrap circuit. (a) Bootstrap capacitor charging, (b) Bootstrap capacitor discharging.

As we remember that MOSFETs Q1 and Q2 should not turned on at the same time (to avoid shoot through or short circuit). Let suppose that for the first cycle MOSFET Q2 is turned on while Q1 is off, the output terminal voltage is pulled to ground (Vout = 0V) and the bootstrap capacitor (Cboot) charges with voltage Vcc

through the bootstrap diode Dboot as shown in Figure 2.27(a). On the next cycle boot strap capacitor discharge, for turn on MOSFET Q1 while MOSFET Q2 is turned off, output voltage Vout is pulled up to battery voltage and the capacitor terminals will be connected between Vout and Vcc which is what makes the high side gate voltage (Vg), as shown on Figure 2.27(b). Diode Dboot will make sure that always gate voltage is greater than source voltage. High side gate voltage measured between gate terminal and ground, equal to the battery voltage (Vbat) plus gate driver power supply voltage (Vcc) minus diode forward voltage VDF.

V_GHS = V_bat+ V_cc− V_DF (2.24)

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Chapter. 3

Design specification

While the decision to design a converter module was made, many factors must be considered in order to meet the target system design requirements and specification.

The steps followed to facilitate designing are the choice of topology and component specification.A list of target converter goals is presented on the following table 3.1.

These began with the targeted battery voltage and current to be converted.

Table 3.1: Design Specification

Specification Value Unit

Specification 2.5 to 4.3 V

Output current 50 A

Switching frequency >10 kHz

Control frequency 50 Hz

3.1 Topology

The topology of our design was based to full bridge converter using N-channel MOS- FET as power switches. External gate resistance was also added in order to limit gate current. As it was not easy to find on chip gate drive compatible to our specification power supply voltage, we had to design discrete one as shown on the following figure.

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Figure 3.1: Design topology with gate drivers

3.2 Components selection

There are many factors to be considered in components’ selection depending on our design requirements and specification, the most important ones are the current rating, the voltage rating and operating frequency.

3.2.1 MOSFET Selection

The key components for converter are the switching elements. There are many types of semiconductor switching devices used in converter design, depending on different factors there are thyristors, IGBT, BJT and MOSFET. Based on our converter design requirement (to convert high current lithium battery of 4V/50 A), we choose MOSFET.

The advantages of MOSFET to be controlled by gate voltage which lead to lower power losses and high switching speed compared to BJT that require high base current to turn on, with a relatively slow turn off. Also, BJTs are highly susceptible to thermal runway, due to their negative temperature coefficient compared to the MOSFETs which has positive one.

We did also the comparison between MOSFET and IGBT. IGBT was developed to be able to compensate the problem of parasitic diode and BJT on MOSFET due to its physical structure. MOSFET’s parasitic body drain diode can be used as freewheeling diode in inductive load, however it increases the switching time, consequently MOSFETs have a noticeable switching loss and cannot work with very high voltage.

IGBTs have combined features of MOSFET and BJT which help them to be able to work on high voltage. IGBTs are voltage-controlled devices with large current tail. IGBTs has no body drain diode and have negative temperature coefficient. To improve the switching off delay of IGBTs, addition ultrafast recovery diode is used.

The following table show the resume comparison of power transistors.

28 CHAPTER 3. DESIGN SPECIFICATION

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Table 3.2: Power transistors comparison [14]

Features BJT MOSFET IGBT

Driver Method Current Voltage Voltage

Driver circuits complex Simple simple

Input

impedance Low High High

Switching speed Slow µs Fast ns Middle

Operating

Frequency Low (less than

100 kHz) Fast (less than

10 MHz) Middle

S.A.O (Safe-

Operating-Area) Narrow Wide Wide

Saturation

voltage Low High Low

As MOSFET also has two channel types known as n-channel and p-channel MOS- FET. After some quick comparison between them, we decided to use n-channel MOSF- FETs for their lower loss, because n-channel MOSFETs have much lower On-resistance (Rdson) at lower cost than p-channel MOSFETs, although gate driving for high-side MOSFETs is more complex. Another disadvantage of p-channel MOSFETs, they are slower to switch on and off, this increase dynamic losses and complicate shoot-through protection.

After deciding to use N-channel MOSFET, we looked also on the manufacture standard and availability on the market. We chose LiU Power MOSFET as power MOSFETs, BSD235C for the first stage of gate driver (level shifter) becaus of its lowest input capacitance, and IRF7317 for the second stage of gate drivers (Buffer) design.

3.2.2 Passive components selection

Passive components have been also chosen carefully, to limit power loss of our system by reaching the optimum time response for the converter. Resistances are used to limit the current flow through the semi-conductor devices where required. We have also used the capacitors, one capacitor was used as bootstrap for the high side power MOSFET driver, the remaining capacitor are used as decoupling capacitors to limit voltage spikes, caused by dI/dt through the parasitic inductance in circuit. To choose passive devices, method of iteration was used and simulation until we reach to the optimum results.

CHAPTER 3. DESIGN SPECIFICATION 29

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Table 3.3: Components number and rating of used devices

Note:

• Decoupling capacitors are 6 in parallel with rating of 1000µ/6.3V each.

• Rx stand for gate resistance (Rg), which is the optimum gate resistance to limit the ringing (damped oscillator) which cause the oscillation and EMI noise at the output. It is optimum because when it is high, switching time becomes longer and when it is lower, ringing become high.

• Ry stand for addition resistance between complementary MOSFET (PMOS- NMOS combination) of the gate driver, to limit turn-on and turn-off current as well as the shoot through current when PMOS and NMOS conduct simultaneously during the transition of the gate voltage.

• Rz stand for optimum turn off gate resistance Rgoff to limit turnoff di/dt and dv/dt of the MOSFET when antiparallel diode shunt gate resistance Rg to reduce turn-off delay time.

30 CHAPTER 3. DESIGN SPECIFICATION

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Chapter. 4

Design Simulation

Any design that meet the requirements theoretically is needed to be tested to make sure that the design can works for real life situation. In this chapter the chosen semiconductor parameter should be used in designed converter, then converter was drawn and simulated using Simulink to ensure the design is feasible. The result obtained from simulations for different conditions are discussed and checked. Switching speed and losses are measured and calculated.

4.1 Simulink

The software chosen for the simulation is Simulink, which is add-on software package to MATLAB, a way to model, simulate and analyse the dynamic systems. It supports both the linear and non-linear systems. A graphical user interface (GUI) is provided by Simulink, in which the models are represented by block diagrams and the opera- tions are click and drag mouse based. The models can be created easily by selecting the specific components of the circuit from Simulink library. Components in model can be customized to the existing component on the market for simulation of their behaviour. Simulink library consists also of sources, sinks and several blocks which result in mathematical outputs, and oscilloscope for display and measure the results graphically.

4.1.1 The designed model using Simulink

As the full bridge converter has two legs with similar components and behaviors, we analyze the Simulink model of half bridge converter as shown in figure 4.1.

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Figure 4.1: Simulink model of Half-bridge converter

Half-bridge simulation model is composed with two mains blocks known as PWM block and another block of containing half bridge and gate control driver. It also contains battery voltage source, constant current source to maintain current constant (to maintain current value independent of the voltage applied on circuit’s terminals), and scope for displaying simulation graphs. Parasitic components such as inductance of estimated value of 2 nH from wiring, and the compensation capacitor to limit back emf effect or ^di_dt have been also added to the model.

Inside the half-bridge converter block, high side, and lower side power MOSFETs are connected with their respect gate drivers and also some measurement blocks are placed in order to measure current across any specific section and voltage between across specific node.

Figure 4.2: Half bridge converter with gate driver

4.1.2 Gate control Driver

The designed converter was for very low voltage (3.7V), so that we were not be able to find the suitable on chip gate drive meeting that lower voltage. For that reason, discrete gate drive was designed and combined with external gate resistances Rgon and R_{gof f}. As shown on the figure 4.3 from Simulink subsystem of gate control driver, a complementary pair of MOSFETs have been combined to build a push-pull pair of transistors to increase the drive strength significantly.

32 CHAPTER 4. DESIGN SIMULATION

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Figure 4.3: Simulink model for MOSFET gate control driver block

Two stage of amplification of the driver signal have been combined, where the first stage is for voltage level shifting, and the last one is buffer for current magnification.

Addition components have been incorporated for the connection of the gate drive to the power MOSFET. Addition diode in parallel with gate resistance have been used to allow lower impedance turn-off path and high impedance turn-on path, this combination helps to speed up turn-off delay trough the lower gate resistance path and reducing voltage overshoots.We used two diodes in series so that the low resistance path is interrupted when the gate voltage reaches the threshold to ensure current turn- off dI/dt is not too high. Then another MOSFET known as Miller clamping transistor have been added to protect the MOSFET from high dV/dt feeding current to the gate through the Miller capacitor (CGD) and ensuring a strong clamping force while holding the power MOSFET off.

Our discrete gate driver has the following power supply characteristics.

Table 4.1: Gate driver power supply characteristics.

Specification Value Unit

Minimum power supply 1.4 V

Recommended operation Power supply

2 V

Maximum power supply 20 (Cf. CMOS BSC235C Data

sheet) V

Note: The minimum voltage is the lowest gate supply voltage to switch on the circuits but it takes the long time to charge the input gate capacitance of the power MOSFET for the first time.

CHAPTER 4. DESIGN SIMULATION 33

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Turn-on process

The switch on transition start by applying a positive pulse on the gate driver. This positive pulse will turn on MOSFET QG2 and QG3, which will allow the current to pass through the on-gate resistance composed by RGon in series with RGon2 and charging the input capacitance off power MOSFET.

Turn-off process

The process of turn-off starts by applying a 0 logic PWM on the gate driver. This action will turn on MOSFETs QG1 and QG4. As MOSFET QG4 is on, it will allow the cathode of diode D1 to be forward biased which will short circuits the resistance R_Gon2 to ensure that the power MOSFET will turning off via a small impedance. This will allow the rapid discharge of the input gate capacitance of power MOSFET. The Miller clamping MOSFET Qcl is turned on a short interval later than QG4. It gives rapid discharge down to a level just above the gate threshold to reduce turn-off delay.

The last part of turn-off when current is reduced, is done with higher gate resistance (Rgon2+Rgoff1) since the diode does not conduct. This ensures limitation of dI/dt at turn-off. Clamping MOSFET also forcing the gate terminal and source terminal of the power MOSFET to be at the same potential and protect it against dV/dt induced turn-on.

4.1.3 Pulse width modulation

PWM is also created in Simulink, using comparators, logic blocks and signal source blocks for tringle and sinusoidal signal as shown on the figure bellow.

Figure 4.4: Simulink model PWM block

In the PWM block, triangle signal with peak amplitude of +/- 1 and switching frequency (fsw) is compared to a sine wave signal with peak amplitude of 0.8 V and frequency of 50 Hz. The output signals of the PWM are used to control gates of the powers MOSFETs.

34 CHAPTER 4. DESIGN SIMULATION

Linköpingsuniversitet58183LinköpingSverige Date:2021-02-22 Examiner: LarsEriksson Supervisor: TomasUnoJonsson BillySecondNIYINZI ConvertermoduleforhighcurrentLithiumbattery LinköpingsuniversitetInstitutionenförsystemteknik(ISY)Master’sProgrammeinElectroni

Master’s Programme in Electronics Engineering Master’s thesis 30 hp

2020

LiTH-ISY-EX–20/5355–SE

Converter module for high current Lithium battery

Billy Second NIYINZI Supervisor: Tomas Uno Jonsson

Examiner: Lars Eriksson Date: 2021-02-22

Linköpings universitet 581 83 Linköping

Sverige

CONVERTER MODULE FOR HIGH CURRENT LITHIUM BATTERY

Billy Second Niyinzi

2021-02-22

Abstract

Acknowledgements

Contents

List of Figures

List of Tables

List of Acronyms and Abbreviations

Chapter. 1

Introduction

1.1 Motivation

1.2 Aims (Purpose)

1.3 Methodology (Scope)

Chapter. 2

Theoretical background

2.1 Converter

2.1.1 Full bridge converter

2.1.2 H-Bridge converter application

2.1.3 Modular Multilevel Converter MMC

2.2 Power MOSFET

2.2.1 Basic structure of power MOSFET

2.2.2 Working principle

2.2.3 Characteristics of MOSFET

2.2.4 MOSFET as electronics switch

2.2.5 Switching properties of the MOSFET

2.2.6 Overshoot voltage

2.2.7 Blanking time

2.2.8 MOSFET power losses

2.2.9 MOSFET thermal behavior

2.3 Pulse width modulation (PWM)

2.3.1 Bipolar PWM method

2.3.2 Unipolar PWM

2.4 Gate control driver

2.4.1 Lower-side gate driver

2.4.2 High-side gate driver

Chapter. 3

Design specification

3.1 Topology

3.2 Components selection

3.2.1 MOSFET Selection

3.2.2 Passive components selection

Chapter. 4

Design Simulation

4.1 Simulink

4.1.1 The designed model using Simulink

4.1.2 Gate control Driver

4.1.3 Pulse width modulation